Pulsed level shift and inverter circuits for gan devices

ABSTRACT

GaN-based half bridge power conversion circuits employ control, support and logic functions that are monolithically integrated on the same devices as the power transistors. In some embodiments a low side GaN device communicates through one or more level shift circuits with a high side GaN device. Both the high side and the low side devices may have one or more integrated control, support and logic functions. Some devices employ electro-static discharge circuits and features formed within the GaN-based devices to improve the reliability and performance of the half bridge power conversion circuits.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. provisional patent applicationSer. No. 62/051,160, for “HYBRID HALF-BRIDGE DRIVER USING GAN ANDSILICON DEVICES” filed on Sep. 16, 2014 and to U.S. provisional patentapplication Ser. No. 62/127,725, for “HALF BRIDGE POWER CONVERSIONCIRCUITS USING GAN AND SILICON DEVICES” filed on Mar. 3, 2015 which arehereby incorporated by reference in their entirety for all purposes.

FIELD

The present invention relates generally to power conversion circuits andin particular to power conversion circuits utilizing one or moreGaN-based semiconductor devices.

BACKGROUND

Electronic devices such as computers, servers and televisions, amongothers, employ one or more electrical power conversion circuits toconvert one form of electrical energy to another. Some electrical powerconversion circuits convert a high DC voltage to a lower DC voltageusing a circuit topology called a half bridge converter. As manyelectronic devices are sensitive to the size and efficiency of the powerconversion circuit, new half bridge converter circuits and componentsmay be required to meet the needs of new electronic devices.

SUMMARY

In some embodiments a half bridge circuit comprising a low side circuitdisposed on a first GaN device and a high side circuit disposed on asecond GaN device is disclosed. The low side circuit includes a low sideswitch having a low side switch control gate and a low side switchdriver having an output connected to the low side switch control gate.The high side circuit includes a high side switch having a high sidecontrol gate and a high side switch driver having an output connected tothe high side switch control gate.

In some embodiments the half bridge circuit may further include a lowside control circuit coupled to the high side and the low side switchdrivers. In further embodiments a level shifter may be configured tocouple one or more signals from the low side control circuit to the highside switch driver. In other embodiments the high side circuit includesa level shift receiver coupled to the level shifter, and the level shiftreceiver includes a signal modulator that is coupled to the high sideswitch driver.

In some embodiments the half bridge circuit includes a level shifterhaving an inverter comprising a resistor pull up and a pull downtransistor. In other embodiments the half bridge circuit includes one ormore pulse generators and a shoot through protection circuit configuredto prevent simultaneous conduction of high side and low side switches.In further embodiments at least one of the low side switch driver andthe high side switch driver have at least one delay circuit. In yetfurther embodiments the low side circuit includes a startup circuit. Insome embodiments the high side circuit includes a high side controllercoupled to the high side switch driver, and the low side circuitincludes a low side controller coupled to the low side switch driver andthe high side controller. In other embodiments at least one of the lowside circuit and the high side circuit have an ESD clamp circuit.

In some embodiments an electronic power conversion component includes apackage base, a first die secured to the package base and comprising alow side circuit, a second die secured to a package base and comprisinga high side circuit and an electrically insulative mold compoundencapsulating at least a portion of a top surface of the package baseand the first and the second dies. In further embodiments the low sidecircuit may include a low side switch having a low side switch controlgate and a low side switch driver having an output connected to the lowside switch control gate. In yet further embodiments the high sidecircuit may include a high side switch having a high side control gate,and a high side switch driver having an output connected to the highside switch control gate.

In some embodiments the package base includes a leadframe. In otherembodiments the component may include an insulator mounted to theleadframe, where the first die is mounted to the leadframe and thesecond die is mounted to the insulator. In other embodiments the packagebase includes a printed circuit board. In further embodiments at leastone of the first and the second die comprise GaN. In yet furtherembodiments the component may have at least one electrical connectionfrom the first die to the second die, formed within the component.

In some embodiments a method of operating a half bridge power conversioncircuit includes operating a low side switch using a low side driver,wherein the low side switch and the low side driver are disposed on afirst GaN device. The method may further include operating a high sideswitch using a high side driver, wherein the high side switch and thehigh side driver are disposed on a second GaN device. In furtherembodiments the method may include controlling the low side driver andthe high side driver with a control circuit that transmits on and offsignals to the low side and the high side drivers. In some embodimentsthe method may comprise transmitting control signals from a low sidecontrol circuit through a level shifter to the high side switch driver.In further embodiments the control signals may be received by a levelshift receiver that modulates the control signals and transmits them tothe high side switch driver.

In some embodiments a level shift circuit comprising a first GaN-basedinverter circuit is disclosed. The inverter circuit may include a firstinput terminal, a first output terminal and a first inversion circuitcoupled between the first input and the first output terminals. Theinverter circuit may be configured to receive a first input logic signalat the first input terminal and in response, provide a first invertedoutput logic signal at the first output terminal. In other embodimentsthe first input and the first inverted output logic signals can bereferenced to different voltage potentials.

In some embodiments the first inversion circuit is configured to becapable of operating with the first inverted output logic signalreferenced to a voltage that is more than 20 volts higher than areference voltage for the first input logic signal. In other embodimentsthe first inversion circuit comprises a first GaN-based enhancement-modetransistor having a gate coupled to the first input terminal, a draincoupled to the first output terminal, and a source coupled to a ground.In further embodiments the first inversion circuit further comprises acurrent sink device coupled between the source and the ground.

In some embodiments, the first inversion circuit further comprises apull up device coupled between the drain and a floating power supply. Inother embodiments the first input logic signal controls on and offtransitions of a high side gate. In one embodiment there is at least onelogic gate configured to prevent simultaneous conduction of high and lowside transistors. In other embodiments the first inverted output logicsignal at the first output terminal is transmitted to a receiver circuitcomprising a driver circuit configured to deliver a voltage above afloating power supply.

In some embodiments the level shift circuit includes an active pull-updevice configured to shorten a time required to reset the first invertedoutput logic signal to a positive state when the first input logicsignal changes from a high state to a low state. In some embodimentsthere may be a first capacitance between the first output terminal and afloating voltage and a second capacitance between the first outputterminal and ground, wherein the first capacitance is greater than thesecond capacitance. In other embodiments an overvoltage condition on thefirst output terminal is prevented by a clamp. In one embodiment afloating supply voltage signal is measured, and in response, a supplyvoltage logic signal is generated and combined with the first invertedoutput logic signal. In other embodiments the supply voltage logicsignal is coupled with a hysteretic inverter.

In some embodiments the level shift circuit further comprises a secondGaN-based inverter circuit having a second input terminal and a secondoutput terminal. A second inversion circuit may be coupled between thesecond input and the second output terminals and configured to receive asecond input logic signal at the second input terminal and in response,provide a second inverted output logic signal at the second outputterminal. In further embodiments the second inversion circuit comprisesa second GaN-based enhancement-mode transistor having a gate coupled tothe second input terminal, a drain coupled to the second outputterminal, and a source coupled to ground. In yet further embodiments thefirst input logic signal is received from a level shift driver and thesecond input logic signal is received from a pulse generator. In oneembodiment the second inverted output logic signal is transmitted to acircuit configured to prevent a change in the first inverted outputlogic signal.

In some embodiments an electronic power conversion component comprisinga package base and one or more GaN-based dies secured to the packagebase is disclosed. The one or more GaN-based dies may include a firstGaN-based inverter circuit comprising a first input terminal and a firstoutput terminal. A first inversion circuit may be coupled between thefirst input and the first output terminals and configured to receive afirst input logic signal at the first input terminal and in response,provide a first inverted output logic signal at the first outputterminal. In further embodiments the first input and the first invertedoutput logic signals can be referenced to different voltage potentials.

In some embodiments the first inversion circuit is configured to becapable of operating with the first inverted output logic signalreferenced to a voltage that is more than 20 volts higher than areference voltage for the first input logic signal. In other embodimentsthe first inversion circuit comprises a first GaN-based enhancement-modetransistor having a gate coupled to the first input terminal, a draincoupled to the first output terminal, and a source coupled to a ground.

In some embodiments a method of operating GaN-based level shift circuitis disclosed. The method may include transmitting a first input logicsignal to a first input terminal and in response, a first inversioncircuit providing an inverted first output logic signal on a firstoutput terminal to control a gate of a power transistor. In oneembodiment the first input logic signal and the inverted first outputlogic signals are referenced to different voltages.

In some embodiments a level shift circuit comprising a first invertercircuit and a second inverter circuit is disclosed. The first invertercircuit may comprise a first input terminal, a first output terminal anda first GaN-based enhancement-mode transistor. The first GaN-basedenhancement-mode transistor has a gate coupled to the first inputterminal, a drain coupled to the first output terminal and a sourcecoupled to a ground. The second inverter circuit may have a second inputterminal, a second output terminal and a second GaN-basedenhancement-mode transistor. The second GaN-based enhancement-modetransistor has a gate coupled to the second input terminal, a draincoupled to the second output terminal and a source coupled to theground.

In some embodiments the first and the second input terminals arereferenced to a first voltage that is a ground, and the first and thesecond output terminals are referenced to a second voltage at adifferent potential than ground. In one embodiment the first inversioncircuit further comprises a pull up device coupled between the drain anda floating power supply. In other embodiments a first capacitance iscoupled between the first output terminal and a floating voltage and asecond capacitance is coupled between the first output terminal andground, wherein the first capacitance is greater than the secondcapacitance.

In some embodiments an overvoltage condition on the first outputterminal is prevented by a clamp. In further embodiments the firstinverter circuit input terminal is configured to receive a first pulsedinput signal from a first pulse generator and the second invertercircuit input terminal is configured to receive a second pulsed inputsignal from a second pulse generator. In one embodiment at least one ofthe first pulse generator and the second pulse generators are configuredto receive input pulses in a range of 2 nanoseconds to 20 microsecondsand to transmit pulses of substantially constant duration within therange. In further embodiments at least one of the first pulse generatorand the second pulse generators comprise at least one combinatoriallogic function.

In some embodiments the input signals from the first and the secondpulse generators correspond to on and off transitions of a pulse-widthmodulated (PWM) signal controlling a gate of a high side transistor. Infurther embodiments the level shift circuit further comprises a latchingstorage logic circuit configured to change state in response to a firstpulsed input signal from the first pulse generator and to change statein response to a second pulsed input signal from the second pulsegenerator. In one embodiment the first and the second pulsed inputsignals from the first and the second pulse generators, respectively,correspond to on and off transitions of a PWM signal to control the gateof a high side transistor. In yet further embodiments at least one ofthe first and the second pulse generators are coupled with one or morelogic gates. In other embodiments the level shift circuit is furtherconfigured to generate a logical combination of at least one PWM signaland at least one pulse generator output signal wherein the logicalcombination is used to prevent simultaneous conduction of a high sideand a low side switch.

In some embodiments an on level shift pulse can be shortened by an offinput pulse to enable an on time of less than 50 nanoseconds on a highside switch. In one embodiment an off level shift pulse can be shortenedby an on input pulse to enable an off time of less than 50 nanosecondson a high side switch. In other embodiments the first output terminal iscoupled to a circuit configured to charge a state storage capacitorreferenced to the second voltage. In further embodiments the secondoutput terminal is coupled to a circuit configured to discharge a statestorage capacitor that is referenced to the second voltage. In yetfurther embodiments an output signal from one of the first or the secondoutput terminals prevents a dv/dt induced change in a signal from theother output terminal.

In some embodiments an electronic power conversion component includes apackage base and one or more GaN-based dies secured to the package base.The one or more GaN-based dies include a first inverter circuitcomprising a first input terminal and a first output terminal. A firstGaN-based enhancement-mode transistor has a gate coupled to the firstinput terminal, a drain coupled to the first output terminal, and asource coupled to a ground. The one or more GaN-based dies include asecond inverter circuit comprising a second input terminal and a secondoutput terminal. A second GaN-based enhancement-mode transistor has agate coupled to the second input terminal, a drain coupled to the secondoutput terminal, and a source coupled to the ground.

In some embodiments a method of operating GaN-based level shift circuitis disclosed. The method includes generating a first pulse with a firstpulse generator, the first pulse operating a first inverter circuitconfigured to change a state of a state storage device. The methodfurther includes generating a second pulse with a second pulsegenerator, the second pulse operating a second inverter circuitconfigured to change a state of the state storage device.

In some embodiments a charging circuit comprising a GaN-basedsemiconductor circuit configured to allow unidirectional current flowfrom a ground referenced power supply to a floating power supplyterminal is disclosed. In one embodiment the semiconductor circuit isconfigured to be capable of operating with the floating power supplyterminal at a voltage that is 20 volts or greater than a voltage of theground referenced power supply. In further embodiments the semiconductorcircuit comprises at least one of: a schottky diode, an enhancement-modetransistor or a depletion-mode transistor. In yet further embodimentsthe semiconductor circuit comprises an enhancement-mode transistor thatincludes a gate and a source connected to a common voltage potential.

In some embodiments the drain of the enhancement-mode transistor isconnected to the floating power supply terminal. In one embodiment thesemiconductor circuit comprises an enhancement transistor that includesa gate that is controlled by a gate drive circuit. In other embodimentsthe drain of the enhancement-mode transistor is connected to thefloating power supply terminal. In further embodiments theenhancement-mode transistor includes a drain that is connected to asource of a depletion-mode transistor and a drain of the depletion-modetransistor is connected to the floating power supply terminal. In yetfurther embodiments a gate of the depletion-mode transistor is connectedto the ground referenced power supply.

In some embodiments a gate of the depletion-mode transistor is connectedto ground. In one embodiment the semiconductor circuit is used inconjunction with a half bridge circuit comprising a low side GaN-basedtransistor having a low side transistor control gate configured toreceive a low side gate signal from a ground referenced gate drivecircuit, and a high side GaN-based transistor having a high sidetransistor control gate configured to receive a high side gate signalfrom a gate drive circuit that is referenced to a second floating powersupply terminal. In further embodiments the second floating power supplyterminal is a switch node of the half bridge circuit. In yet furtherembodiments a capacitor is connected between the floating power supplyterminal and the second floating power supply terminal.

In some embodiments the semiconductor circuit comprises anenhancement-mode transistor including a gate that is controlled by agate drive circuit and the gate drive circuit is configured such that itprovides an output voltage that is in phase with the low side gatesignal. In further embodiments a delay circuit is configured to turn onthe enhancement-mode transistor after the low side GaN-based transistorturns on. In yet further embodiments a delay circuit is configured toturn off the enhancement-mode transistor before the low side GaN-basedtransistor turns off.

In some embodiments an electronic power conversion component includes apackage base and one or more GaN-based dies secured to the package baseincluding a charging circuit. In further embodiments the chargingcircuit comprises a GaN-based semiconductor circuit configured to allowunidirectional current flow from a ground referenced power supply to afloating power supply terminal. In some embodiments the semiconductorcircuit comprises at least one of a: a schottky diode, anenhancement-mode transistor and a depletion-mode transistor. In furtherembodiments the semiconductor circuit includes an enhancement-modetransistor having a drain that is connected to a source of adepletion-mode transistor, where a drain of the depletion-modetransistor is connected to the floating power supply terminal.

In some embodiments a method of operating GaN-based charging circuit isdisclosed. The method includes supplying power with a ground referencedpower supply to a first terminal of a GaN-based semiconductor circuit.Current is allowed to flow through the GaN-based semiconductor circuitonly in a direction from the first terminal to a second terminal, andthe second terminal is a floating power supply.

In some embodiments a power supply circuit comprising a GaN-baseddepletion-mode transistor used as one of a voltage-limited voltagesource or a voltage-limited current source is disclosed. In oneembodiment the depletion-mode transistor is used in a reference circuitto set a reference voltage and includes a first drain coupled to a powersource and a first source coupled to a first node. In another embodimenta first gate of the depletion-mode transistor is connected to ground. Inanother embodiment a first gate of the depletion-mode transistor isformed by a metal layer disposed over a passivation layer. In furtherembodiments the depletion-mode transistor is disposed on a GaN-basedpower integrated circuit device.

In some embodiments the power supply circuit further comprises aplurality of series connected circuit elements coupled between the firstnode and a second node, and one or more intermediate nodes disposedbetween each of the plurality of series connected circuit elements. Inone embodiment the power supply circuit further comprises a GaN-basedreference voltage transistor having a second gate connected to one ofthe one or more intermediate nodes, and a second source configured todeliver power to a circuit and a second drain connected to a powersource. In further embodiments the GaN-based reference voltagetransistor includes one or more diodes or diode-connected transistorsdisposed between the second gate and the second source, configured asgate overvoltage protection devices.

In some embodiments the power supply circuit may further comprise adisable circuit configured to prevent the second source from deliveringpower to a circuit. In one embodiment the reference voltage transistoris a GaN-based enhancement-mode transistor. In another embodiment thepower supply circuit is configured to be a ground referenced powersupply in a half bridge circuit. In further embodiments the second nodeis connected to ground. In another embodiment a capacitor is connectedbetween the first node and the second node. In yet further embodimentsat least one of the first node and the second node are connected to acapacitor. In other embodiments a diode or a diode-connected transistoris coupled between the first node and a circuit configured to deliverpower.

In some embodiments the power source comprises a floating voltage. Inanother embodiment the reference circuit is configured to supply poweronly when the power source is within a predetermined range. In furtherembodiments the power source has a constantly varying voltage. In yetfurther embodiments the power source is an AC line voltage. In otherembodiments the power supply circuit further comprises a thirdenhancement-mode transistor having a third gate, a third source and athird drain, and a fourth enhancement-mode transistor having a fourthgate, a fourth source and a fourth drain. The third and the fourthsources are coupled to a third node, the third gate and fourth gates arecoupled together, the third drain is coupled to the second node and thefourth drain is coupled to a reference current sink terminal. In someembodiments the power supply circuit further comprises a comparatorcircuit coupled to a ground referenced power supply and the referencecurrent sink terminal.

In some embodiments an electronic power conversion component comprisinga package base having one or more GaN-based dies secured to the packagebase and including a power supply circuit is disclosed. In oneembodiment a GaN-based depletion-mode transistor is used as one of avoltage-limited voltage source or a voltage-limited current source.

In some embodiments a method of operating GaN-based power supply circuitis disclosed. The method includes supplying power to a drain terminal ofa GaN-based depletion-mode device having a first gate connected toground and a first source connected to one or more series connectedcircuit elements including one or more intermediate nodes between eachof the plurality of series connected circuit elements. The methodfurther includes delivering power to one or more circuits from a secondsource of a GaN-based enhancement-mode device having a second gatecoupled to one of the one or more intermediate nodes and a second drainconnected to a power source.

In some embodiments a semiconductor device comprising a level shifttransistor having a ratio of output saturation current (Idsat) to outputcapacitor charge (Qoss) of greater than 1 A/nc is disclosed. In oneembodiment the level shift transistor is GaN-based. In anotherembodiment the level shift transistor has less than 25 pC of outputcharge (Qoss). In further embodiments the level shift transistor isoperated with a pulsed input signal. In yet further embodiments aduration of the pulsed input signal is less than 100 ns. In someembodiments a channel width of the level shift transistor is less than100 microns. In yet further embodiments a drain structure of the levelshift transistor is placed less than 100 microns from a bond pad.

In some embodiments the level shift transistor includes a source ohmiccontact area connected to a source terminal, and the source terminal isconnected to a metal pad that is immediately adjacent to the sourceterminal and is more than 100 times the source ohmic contact area. Inother embodiments the level shift transistor includes a drain ohmiccontact area connected to a drain terminal and the drain terminal isconnected to a metal pad that is immediately adjacent to the drainterminal and is more than 100 times the drain ohmic contact area. Infurther embodiments the level shift transistor comprises a source areaand a drain area and the source area does not encircle the drain area.In yet further embodiments the level shift transistor comprises anactive region having a source area at a first end and a drain area at anopposing end.

In some embodiments a level shift circuit comprising an input referencedto ground and an output referenced to a floating voltage is disclosed.The circuit is configured to be integrated on at least one GaN device.In some embodiments the level shift circuit includes a transistor havingan Idsat to Qoss ratio greater than lA/nc. In other embodiments thelevel shift circuit includes a first capacitance between the output andthe floating voltage, where the first capacitance is configured toprevent a change of output state when the floating voltage changesvoltage potential from ground to a maximum allowed voltage. In otherembodiments the level shift circuit comprises an electrically conductivecircuit element coupled between a source of a level shift transistor andground.

In some embodiments the level shift circuit comprises an electricallyconductive circuit element coupled between a drain of a level shifttransistor drain and a positive side of a power source that isreferenced to the floating voltage. In further embodiments the levelshift circuit includes a first circuit portion disposed on a first GaNdevice and a second circuit portion disposed on a second GaN device. Insome embodiments the first circuit portion comprises the output and thesecond circuit portion comprises a receiver circuit, and a bond wireforms an electrical connection between the output and the receivercircuit.

In some embodiments the level shift circuit comprises at least oneoutput terminal bond pad having a conductive shield underneath it thatis referenced to the floating voltage. In other embodiments at least onelevel shift transistor and all ground referenced circuit elements aredisposed on the first GaN device. In one embodiment the level shiftcircuit comprises a low side power switch disposed on the first GaNdevice. In further embodiments the second circuit portion comprises anelectrically conductive circuit element coupled between a drain of alevel shift transistor drain and a positive side of a power source thatis referenced to the floating voltage. In yet further embodiments thelevel shift circuit comprises a high side power switch integrated on thesame device.

In some embodiments a circuit including overvoltage protection isdisclosed. The circuit comprises a first pin and a second pin, and anovervoltage protection circuit comprising a first enhancement-modetransistor disposed on a GaN-based substrate and coupled between thefirst pin and the second pin. In some embodiments the overvoltageprotection circuit does not contain depletion-mode transistors. Infurther embodiments the overvoltage protection circuit comprises asecond enhancement-mode transistor having a source coupled to a gate ofa third enhancement-mode transistor, and an electrically conductiveelement coupled in an electrical path between the source and the secondpin. The electrically conductive element includes one of a resistor, adepletion-mode transistor, a reference current sink or a referencecurrent source.

In some embodiments the overvoltage protection circuit comprises asecond enhancement-mode transistor having a source coupled to a gate ofa third enhancement-mode transistor. An electrically conductive elementis coupled in an electrical path between the source and the gate. Theelectrically conductive element comprises one of a resistor, adepletion-mode transistor, a reference current sink or a referencecurrent source. In further embodiments the first pin is the gate of apower transistor and the second pin is the source of the powertransistor. In one embodiment the overvoltage protection circuit iscoupled between a power supply terminal and ground.

In some embodiments the overvoltage protection circuit is configured toremain in an off state until a voltage potential across the first andthe second pins is above a predetermined voltage level. In furtherembodiments the first enhancement-mode transistor has a first sourcecoupled to the first pin and a first drain coupled to the second pin.The first enhancement-mode transistor is configured to provideovervoltage protection between the first and the second pins. In oneembodiment a first gate of the first enhancement-mode transistor iscoupled to the first source and the first enhancement-mode transistor isconfigured to remain in an off state until it is subjected to anovervoltage pulse. In some embodiments one or more diodes ordiode-connected transistors are connected in series and coupled to agate of the first enhancement-mode transistor, configured to drive thefirst enhancement-mode transistor.

In some embodiments the first enhancement-mode transistor is configuredto conduct a current greater than 500 mA when exposed to an overvoltagepulse. In one embodiment the overvoltage protection circuit comprisessecond and third enhancement-mode transistors. A third source of thethird enhancement-mode transistor is connected to a second gate of thesecond enhancement-mode transistor, and a second source of the secondenhancement-mode transistor is connected to a first gate of the firstenhancement-mode transistor. In further embodiments a disable circuit isconfigured to prevent current flow between the first and the second pinsfor a predetermined dv/dt value occurring at the first or the second pinthat is less than 1 V/ns. In yet further embodiments the disable circuitcomprises a dv/dt detection filter coupled to the first gate of thefirst enhancement-mode transistor.

In some embodiments the dv/dt detection filter comprises at least oneGaN-based logic circuit. In one embodiment the first enhancement-modetransistor is connected in series with a source of a depletion-modetransistor. A drain of the depletion-mode transistor is connected to thefirst pin, and the first source is connected to the second pin. In otherembodiments the overvoltage protection circuit comprises a secondenhancement-mode transistor connected in parallel with the firstenhancement-mode transistor. The overvoltage protection circuit isconfigured to provide symmetric overvoltage protection for the circuitwhen exposed to either positive or negative overvoltage conditions. Infurther embodiments the overvoltage protection circuit comprises asecond enhancement-mode transistor connected in series with the firstenhancement-mode transistor, and the overvoltage protection circuit isconfigured to provide symmetric overvoltage protection for the circuitwhen exposed to either positive or negative overvoltage conditions.

In some embodiments an ESD protection circuit comprising a GaN-basedcircuit having two pins is disclosed. A first enhancement-modetransistor is coupled between the two pins and has a first gate. A dv/dtdetection filter is coupled to the gate and is configured to enablecurrent flow between the two pins when a dv/dt on at least one of thetwo pins is a value greater than 1 V/ns. In some embodiments the ESDprotection circuit further comprises an overvoltage protection circuitthat includes an enhancement-mode-transistor coupled between the twopins and configured to temporarily conduct current between the two pinswhile a voltage potential between the two pins is above a predeterminedlevel.

In some embodiments an electronic power conversion component comprisinga package base and one or more GaN-based dies secured to the packagebase is disclosed. The one or more GaN-based dies include a firstcircuit comprising at least one enhancement-mode transistor, and anovervoltage protection circuit coupled to the first circuit.

In some embodiments a method of operating a GaN-based circuit isdisclosed. The method includes receiving a voltage potential above apredetermined value across two pins of a circuit and turning on aGaN-based enhancement-mode transistor coupled between the two pins. Theenhancement-mode transistor temporarily conducts current between the twopins while the voltage potential is above the predetermined value. Insome embodiments the method further comprises receiving a dv/dt signallarger than 1 V/ns on at least one of the two pins, and in responseturning on a second GaN-based enhancement-mode transistor enablingcurrent to flow between the two pins.

In some embodiments an electronic circuit including a substratecomprising GaN is disclosed. A power switch is formed on the substrateand includes a first control gate and a first source. A drive circuit isformed on the substrate and includes an output coupled to the firstgate. A power supply has a maximum voltage and is coupled to the drivecircuit, where the output can be driven to the maximum voltage. Infurther embodiments the drive circuit is coupled to at least one powersupply and to one input that are both referenced to the first source. Insome embodiments the drive circuit is coupled to exactly one PWM input.In other embodiments the drive circuit includes at least oneenhancement-mode transistor, at least one current conducting element anddoes not include any depletion-mode transistors.

In some embodiments the drive circuit comprises an inverter thatincludes a first enhancement-mode transistor having a second gateconnected to a first input signal, a second source connected to thefirst source, and a second drain. A second enhancement-mode transistorhas a third drain connected to the power supply, a third sourceconnected to the second drain and a third gate connected to a circuitconfigured to generate a voltage higher than the power supply. In oneembodiment a capacitive element moves up and down in voltagesynchronously with the third source and supplies power to the thirdgate. In some embodiments a rectifying element is configured to supplypower to the capacitive element and prevent discharge of the capacitiveelement when a terminal of the capacitive element rises above a voltageon the power supply.

In some embodiments the second enhancement-mode transistor can beswitched on in less than 100 nanoseconds. In one embodiment a thirdenhancement-mode transistor has a fourth gate connected the first inputsignal, a fourth drain connected to the third gate and a fourth sourceconnected to the first source. In other embodiments a current limitingelement is disposed in a current conduction path from the power supplyto the first source. The current conduction path comprises a seriesconnection of a rectifying element, the current limiting element and thethird enhancement-mode transistor. In further embodiments a resistor isdisposed between the first input signal and the control gate. In yetfurther embodiments the drive circuit comprises two inverters connectedserially to form a non-inverting buffer circuit. In one embodiment thedrive circuit comprises at least one buffer circuit. In otherembodiments the drive circuit is coupled with a gate of a fourthenhancement-mode transistor having a fifth drain connected to thecontrol gate and a fifth source connected to the first source. Furtherembodiments include an electrostatic discharge protection circuit.

In some embodiments an electronic component comprising a package basehaving at least one GaN-based die secured to the package base andincluding an electronic circuit is disclosed. A power switch is formedon the at least one GaN based die and includes a first control gate anda first source. A drive circuit is formed on the at least one GaN baseddie and includes an output coupled to the control gate. A power supplyhaving a maximum voltage is coupled to the drive circuit, where theoutput can be driven to the maximum voltage. In one embodiment the drivecircuit is coupled to at least one power supply and to one input thatare referenced to the first source. In another embodiment the drivecircuit is coupled to exactly one PWM input.

In some embodiments the drive circuit further includes at least oneenhancement-mode transistor, at least one current conducting element,and does not include any depletion-mode transistors.

In some embodiments a method of operating GaN-based circuit isdisclosed. The method includes receiving a signal with a drive circuitand processing the signal with the drive circuit. A signal istransmitted to a control gate of a switch and the drive circuit and theswitch are disposed on a unitary GaN substrate. The drive circuitincludes at least one enhancement-mode transistor, at least one currentconducting element and does not include any depletion-mode transistors.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic of a half bridge power conversioncircuit according to an embodiment of the invention;

FIG. 2 is a simplified schematic of the circuits within the low sidecontrol circuit illustrated in FIG. 1;

FIG. 3 is a schematic of the first level shift transistor illustrated inFIG. 1;

FIG. 4 is a schematic of the level shift driver circuit illustrated inFIG. 1;

FIG. 5 is a schematic of the blanking pulse generator circuitillustrated in FIG. 1;

FIG. 6 is an example of waveforms within the blanking pulse generatorillustrated in FIG. 5;

FIG. 7 is a schematic of the bootstrap transistor drive circuitillustrated in FIG. 1;

FIG. 8 is a block diagram for the low side transistor drive circuitillustrated in FIG. 1

FIG. 9 is a schematic of the startup circuit illustrated in FIG. 1;

FIG. 10 is series of diode connected GaN-based enhancement-modetransistors that may be used as a diode clamp in the schematic of FIG.9;

FIG. 11 is a schematic of the UVLO circuit illustrated in FIG. 1;

FIG. 12 is a schematic of the bootstrap capacitor charging circuitillustrated in FIG. 1;

FIG. 13 is a schematic of an alternative bootstrap capacitor chargingcircuit as compared to the circuit illustrated in FIG. 12;

FIG. 14 is a schematic of the high side logic and control circuitillustrated in FIG. 1;

FIG. 15 is a schematic of the first level shift receiver circuitillustrated in FIG. 14;

FIG. 16 is a schematic of the second level shift receiver circuitillustrated in FIG. 14;

FIG. 17 is a schematic of the pull up trigger circuit illustrated inFIG. 14;

FIG. 18 is a schematic of the high side UVLO circuit illustrated in FIG.14;

FIG. 19 is a schematic of the high side transistor driver circuitillustrated in FIG. 14;

FIG. 20 is a schematic of a high side reference voltage generationcircuit illustrated in FIG. 14;

FIG. 21 is a simplified schematic of a half bridge power conversioncircuit according to another embodiment of the invention;

FIG. 22 is a simplified schematic of the circuits within the low sidecontrol circuit illustrated in FIG. 21;

FIG. 23 is a schematic of the first level shift transistor illustratedin FIG. 22;

FIG. 24 is a schematic of the inverter/buffer circuit illustrated inFIG. 22;

FIG. 25 is a schematic of the on pulse generator circuit illustrated inFIG. 22;

FIG. 26 is a schematic of the off pulse generator circuit illustrated inFIG. 22;

FIG. 27 is a schematic of the blanking pulse generator circuitillustrated in FIG. 22;

FIG. 28 is a schematic of the low side transistor drive circuitillustrated in FIG. 22;

FIG. 29 is a simplified schematic of the circuits within the high sidecontrol circuit illustrated in FIG. 21;

FIG. 30 is a schematic of the level shift 1 receiver circuit illustratedin FIG. 29;

FIG. 31 is a schematic of level shift 2 receiver circuit illustrated inFIG. 29;

FIG. 32 is a schematic of the high side UVLO circuit illustrated in FIG.29;

FIG. 33 is a schematic of the high side transistor driver circuitillustrated in FIG. 29;

FIG. 34 is a schematic of an electro-static discharge (ESD) clampcircuit according to an embodiment of the invention;

FIG. 35 is a schematic of an electro-static discharge (ESD) clampcircuit according to an embodiment of the invention;

FIG. 36 is an illustration of a portion of an electronic packageaccording to an embodiment of the invention; and

FIG. 37 is an illustration of the electronic package of FIG. 36.

DETAILED DESCRIPTION

Certain embodiments of the present invention relate to half bridge powerconversion circuits that employ one or more gallium nitride (GaN)devices. While the present invention can be useful for a wide variety ofhalf bridge circuits, some embodiments of the invention are particularlyuseful for half bridge circuits designed to operate at high frequenciesand/or high efficiencies with integrated driver circuits, integratedlevel shift circuits, integrated bootstrap capacitor charging circuits,integrated startup circuits and/or hybrid solutions using GaN andsilicon devices, as described in more detail below.

Half Bridge Circuit #1

Now referring to FIG. 1, in some embodiments circuit 100 may include apair of complementary power transistors (also referred to herein asswitches) that are controlled by one or more control circuits configuredto regulate power delivered to a load. In some embodiments a high sidepower transistor is disposed on a high side device along with a portionof the control circuit and a low side power transistor is disposed on alow side device along with a portion of the control circuit, asdescribed in more detail below.

The integrated half bridge power conversion circuit 100 illustrated inFIG. 1 includes a low side GaN device 103, a high side GaN device 105 aload 107, a bootstrap capacitor 110 and other circuit elements, asillustrated and discussed in more detail below. Some embodiments mayalso have an external controller (not shown in FIG. 1) providing one ormore inputs to circuit 100 to regulate the operation of the circuit.Circuit 100 is for illustrative purposes only and other variants andconfigurations are within the scope of this disclosure.

In one embodiment, low side GaN device 103 may have a GaN-based low sidecircuit 104 that includes a low side power transistor 115 having a lowside control gate 117. Low side circuit 104 may further include anintegrated low side transistor driver 120 having an output 123 connectedto low side transistor control gate 117. In another embodiment high,side GaN device 105 may have a GaN-based high side circuit 106 thatincludes a high side power transistor 125 having a high side controlgate 127. High side circuit 106 may further include an integrated highside transistor driver 130 having an output 133 connected to high sidetransistor control gate 127.

A voltage source 135 (also known as a rail voltage) may be connected toa drain 137 of high side transistor 125, and the high side transistormay be used to control power input into power conversion circuit 100.High side transistor 125 may further have a source 140 that is coupledto a drain 143 of low side transistor 115, forming a switch node 145.Low side transistor 115 may have a source 147 connected to ground. Inone embodiment, low side transistor 115 and high side transistor 125 maybe GaN-based enhancement-mode field effect transistors. In otherembodiments low side transistor 115 and high side transistor 125 may beany other type of device including, but not limited to, GaN-baseddepletion-mode transistors, GaN-based depletion-mode transistorsconnected in series with silicon based enhancement-mode field-effecttransistors having the gate of the depletion-mode transistor connectedto the source of the silicon-based enhancement-mode transistor, siliconcarbide based transistors or silicon-based transistors.

In some embodiments high side device 105 and low side device 103 may bemade from a GaN-based material. In one embodiment the GaN-based materialmay include a layer of GaN on a layer of silicon. In further embodimentsthe GaN based material may include, but not limited to, a layer of GaNon a layer of silicon carbide, sapphire or aluminum nitride. In oneembodiment the GaN based layer may include, but not limited to, acomposite stack of other III nitrides such as aluminum nitride andindium nitride and III nitride alloys such as AlGaN and InGaN. Infurther embodiments, GaN-based low side circuit 104 and GaN-based highside circuit 106 may be disposed on a monolithic GaN-based device. Inother embodiments GaN-based low side circuit 104 may be disposed on afirst GaN-based device and GaN-based high side circuit 106 may bedisposed on a second GaN-based device. In yet further embodimentsGaN-based low side circuit 104 and GaN-based high side circuit 106 maybe disposed on more than two GaN-based devices. In one embodiment,GaN-based low side circuit 104 and GaN-based high side circuit 106 maycontain any number of active or passive circuit elements arranged in anyconfiguration.

Low Side Device

Low side device 103 may include numerous circuits used for the controland operation of the low side device and high side device 105. In someembodiments, low side device 103 may include logic, control and levelshift circuits (low side control circuit) 150 that controls theswitching of low side transistor 115 and high side transistor 125 alongwith other functions, as discussed in more detail below. Low side device103 may also include a startup circuit 155, a bootstrap capacitorcharging circuit 157 and a shield capacitor 160, as also discussed inmore detail below.

Now referring to FIG. 2, the circuits within low side control circuit150 are functionally illustrated. Each circuit within low side controlcircuit 150 is discussed below, and in some cases is shown in moredetail in FIGS. 3-14. In one embodiment the primary function of low sidecontrol circuit 150 may be to receive one or more input signals, such asa PWM signal from a controller, and control the operation of low sidetransistor 115, and high side transistor 125.

In one embodiment, first and a second level shift transistors 203, 205,respectively, may be employed to communicate with high side logic andcontrol circuit 153 (see FIG. 1). In some embodiments, first level shifttransistor 203 may be a high voltage enhancement-mode GaN transistor. Infurther embodiments, first level shift transistor 203 may be similar tolow side transistor 115 (see FIG. 1) and high side transistor 125,except it may be much smaller in size (e.g., first level shifttransistor may be tens of microns in gate width with minimum channellength).

In other embodiments first level shift transistor 203 may experiencehigh voltage and high current at the same time (i.e. the device mayoperate at the high power portion of the device Safe Operating Area) foras long as high side transistor 125 (see FIG. 1) is on. Such conditionsmay cause relatively high power dissipation, thus some embodiments mayinvolve design and device reliability considerations in the design offirst level shift transistor 203, as discussed in more detail below. Infurther embodiments, a first level shift resistor 207 may be added inseries with a source 210 of first level shift transistor 203 to limitgate 213 to source 210 voltage and consequently the maximum currentthrough the first level shift transistor. Other methods may be employedto limit the current through first level shift transistor 203, and arewithin the scope of this disclosure. Drain 215 of first level shifttransistor 203 may be coupled to high side logic and control circuit 153(see FIG. 1), as discussed in more detail below.

In one embodiment, first level shift transistor 203 may comprise aportion of an inverter circuit having a first input and a first outputand configured to receive a first input logic signal at the first inputterminal and in response, provide a first inverted output logic signalat the first output terminal, as discussed in more detail below. Infurther embodiments the first input and the first inverted output logicsignals can be referenced to different voltage potentials. In someembodiments, first level shift resistor 207 may be capable of operatingwith the first inverted output logic signal referenced to a voltage thatis more than 13 volts higher than a reference voltage for the firstinput logic signal. In other embodiments it may be capable of operatingwith the first inverted output logic signal referenced to a voltage thatis more than 20 volts higher than a reference voltage for the firstinput logic signal, while in other embodiments it may be between 80-400volts higher.

In other embodiments, first level shift resistor 207 may be replaced byany form of a current sink. For example, in one embodiment, source 210of first level shift transistor 203 may be connected to a gate to sourceshorted depletion-mode device. In a further embodiment, thedepletion-mode device may be fabricated by replacing theenhancement-mode gate stack with a high voltage field plate metalsuperimposed on top of the field dielectric layers. The thickness of thefield dielectric and the work function of the metal may be used todetermine the pinch-off voltage of the stack.

In other embodiments first level shift resistor 207 may be replaced by acurrent sink. The current sink may use a reference current (Iref) thatmay be generated by startup circuit 155 (illustrated in FIG. 1 anddiscussed in more detail below). Both the depletion-mode transistor andcurrent sink embodiments may result in a significant device areareduction compared to the resistor embodiment (i.e., because arelatively small depletion-mode transistor would suffice and Iref isalready available from startup circuit 155).

Second level shift transistor 205 may be designed similar to first levelshift transistor 203 (e.g., in terms of voltage capability, currenthandling capability, thermal resistance, etc.). Second level shifttransistor 205 may also be built with either an active current sink or aresistor, similar to first level shift transistor 203. In one embodimentthe primary difference with second level shift transistor 205 may be inits operation. In some embodiments the primary purpose of second levelshift transistor 205 may be to prevent false triggering of high sidetransistor 125 (see FIG. 1) when low side transistor 115 turns off.

In one embodiment, for example, false triggering can occur in a boostoperation when low side transistor 115 turn off results in the loadcurrent flowing through high side transistor 125 while the transistor isoperating in the third quadrant with its gate shorted to its source(i.e., in synchronous rectification mode). This condition may introducea dv/dt condition at switch node (Vsw) 145 since the switch node was ata voltage close to ground when low side transistor 115 was on and thentransitions to rail voltage 135 over a relatively short time period. Theresultant parasitic C*dv/dt current (i.e., where C=Coss of first levelshift transistor 203 plus any other capacitance to ground) can causefirst level shift node 305 (see FIG. 3) to get pulled low which willthen turn on high side transistor 125. In some embodiments thiscondition may not be desirable because there may be no dead timecontrol, and shoot through may occur from high side transistor 125 andlow side transistor 115 being in a conductive state simultaneously.

FIG. 3 illustrates one embodiment showing how first level shifttransistor 203 may be electrically coupled to high side device 105.First level shift transistor 203, located on low side device 103, isillustrated along with a pull up resistor 303 that may be located onhigh side device 105 (see FIG. 1). In some embodiments, first levelshift transistor 203 may operate as a pull down transistor in a resistorpull up inverter.

In further embodiments, when level shift driver circuit 217 (see FIG. 2)supplies a high gate signal (L1_DR) to first level shift transistor 203,a first level shift node 305 gets pulled low which is inverted by highside logic and control circuit 153 (see FIG. 1). The inverted signalappears as a high state signal that turns on high side transistor 137(see FIG. 1) which then pulls the voltage at switch node (Vsw) 145 closeto rail voltage 135.

Conversely, when level shift driver circuit 217 (see FIG. 2) supplies alow gate signal to first level shift transistor 203, a first level shiftnode 305 gets pulled to a high logic state which is inverted by highside logic and control circuit 153 (see FIG. 1). The inverted signalappears as a low logic state signal that turns off high side transistor125. This scheme may result in a non-inverted gate signal to high sidetransistor 125. In further embodiments, first level shift transistor 203may be designed large enough to be able to pull down on first levelshift node 305, but not so large that its drain to source and drain tosubstrate (i.e., the semiconductor substrate) capacitances induce falsetriggering of high side logic and control circuit 153.

In some embodiments pull up resistor 303 may instead be anenhancement-mode transistor, a depletion-mode transistor or a referencecurrent source element. In further embodiments pull up resistor 303 maybe coupled between the drain and the positive terminal of a floatingsupply (e.g., a bootstrap capacitor, discussed in more detail below)that is referenced to a different voltage rail than ground. In yetfurther embodiments there may be a first capacitance between the firstoutput terminal (LS_NODE) 305 and switch node (Vsw) 145 (see FIG. 1) anda second capacitance between the first output terminal and ground, wherethe first capacitance is greater than the second capacitance. The firstcapacitance may be designed such that in response to a high dv/dt signalat switch node (Vsw) 145 (see FIG. 1), a large portion of the C*dv/dtcurrent is allowed to conduct through the first capacitance ensuringthat the voltage at first output terminal 305 tracks the voltage at theswitch node (Vsw). In some embodiments shield capacitor 160 (see FIG. 1)may be designed to act as the first capacitor as described above. Infurther embodiments shield capacitor 160 (see FIG. 1) may be used tocreate capacitance between first output terminal 305 and switch node(Vsw) 145 (see FIG. 1) in half bridge power conversion circuit 100. Inyet further embodiments, shield capacitor 160 (see FIG. 1) may also beused to minimize a capacitance between first output terminal 305 andsubstrate (i.e., the semiconductor substrate). More specifically, insome embodiments shield capacitor 160 may be created by adding aconductive shield layer to the device and coupling the layer to switchnode (Vsw) 145. This structure may effectively create two capacitors.One capacitor is coupled between output terminal 305 and switch node(Vsw) 145, and the other is coupled between the switch node and thesubstrate. The capacitance between output terminal 305 and the substrateis thereby practically eliminated. In further embodiments shieldcapacitor 160 (see FIG. 1) may be constructed on the low side chip 103.

Logic, control and level shifting circuit 150 (see FIG. 2) may haveother functions and circuits such as, but not limited to, a level shiftdriver circuit 217, a low side transistor drive circuit 120, a blankingpulse generator 223, a bootstrap transistor drive circuit 225 and anunder voltage lock out (UVLO) circuit 227, as explained in separatefigures with more detail below.

Now referring to FIG. 4, level shift driver circuit 217 is shown ingreater detail. In one embodiment level shift driver circuit 217 mayinclude a first inverter 405 and a second inverter 410 in a sequentialchain. In further embodiments, since level shift driver circuit 217 maybe driving a small gate width first level shift transistor 203, theremay be no need for a buffer stage.

In one embodiment, level shift driver circuit 217 is driven directly bythe pulse-width modulated high side signal (PWM_HS) from the controller(not shown). In some embodiments the (PWM_HS) signal may be supplied byan external control circuit. In one embodiment the external controlcircuit may be an external controller that is in the same package withhigh side device 105, low side device 103, both devices, or packaged onits own. In further embodiments, level shift driver circuit 217 may alsoinclude logic that controls when the level shift driver circuitcommunicates with first level shift transistor 203 (see FIG. 3). In oneembodiment an optional low side under voltage lock out signal (LS_UVLO)may be generated by an under voltage lock out circuit within level shiftdriver circuit 217. The low side under voltage lock out circuit can beused to turn off level shift driver circuit 217 if either (Vcc) or (Vdd)for the low side (Vdd_LS) go below a certain reference voltage, or afraction of the reference voltage.

In further embodiments level shift driver circuit 217 may generate ashoot through protection signal for the low side transistor (STP_LS)that is used to prevent shoot through arising from overlapping gatesignals on low side transistor 115 and high side transistor 125. Thefunction of the (STP_LS) signal may be to ensure that low side drivercircuit 120 (see FIG. 2) only communicates with the gate terminal of thelow side transistor 115 when the gate signal to high side transistor 125is low. In other embodiments, the output of first inverter 405 may beused to generate the shoot through protection signal (STP_LS) for thelow side transistor 115.

In further embodiments, logic for UVLO and shoot-through protection mayimplemented by adding a multiple input NAND gate to first inverter 405,where the inputs to the NAND gate are the (PWM_HS), (LS_UVLO) and(STP_HS) signals. In yet further embodiments, first inverter 405 mayonly respond to the (PWM_HS) signal if both (STP_HS) and (LS_UVLO)signals are high. In further embodiments, the STP_HS signal may begenerated from the low side gate driver block 120, as explained inseparate figures with more detail.

Now referring to FIG. 5, blanking pulse generator 223 may be used togenerate a pulse signal that corresponds to the turn off transient oflow side transistor 115. This pulse signal may then turn on second levelshift transistor 205 for the duration of the pulse, which triggers acontrol circuit on high side device 105 (see FIG. 1) to prevent falsepull down of first level shift node 305 voltage.

FIG. 5 illustrates a schematic of one embodiment of blanking pulsegenerator 223. In some embodiments a low side transistor 115 gate signal(LS_GATE) is fed as an input to blanking pulse generator 223. The(LS_GATE) signal is inverted by a first stage inverter 505, then sentthrough an RC pulse generator 510 to generate a positive pulse. In someembodiments an inverted signal may be needed because the pulsecorresponds to the falling edge of the (LS_GATE) signal. A capacitor 515in RC pulse generator 510 circuit may be used as a high pass filterallowing the dv/dt at its input to appear across resistor 520. Once thedv/dt vanishes at the input to the RC pulse generator 510, capacitor 515may charge slowly through resistor 520, resulting in a slow decayingvoltage waveform across the resistor. The pulse may then be sent througha second inverter 525, a third inverter 530 and a buffer 535 to generatea square wave pulse for the blanking pulse (B_PULSE) signal. Theduration of the pulse may be determined by the value of capacitor 515and resistor 520 in RC pulse generator 510. In some embodiments,capacitor 515 may be constructed using a drain to source shortedenhancement-mode GaN transistor.

Now referring to FIG. 6, example waveforms 600 within blanking pulsegenerator 223 are illustrated for one embodiment. Trace 605 shows afalling edge of the low side gate pulse (LS_GATE). Trace 610 shows therising edge of first stage inverter 505 output. Trace 615 shows theoutput of RC pulse generator 510 and trace 620 shows the resultingblanking pulse (B_PULSE) signal that is an output of blanking pulsegenerator 223.

Now referring to FIG. 7, bootstrap transistor drive circuit 225 isillustrated in greater detail. Bootstrap transistor drive circuit 225includes inverter 730, first buffer 735 and second buffer 745. Bootstraptransistor drive circuit 225 may receive the (BOOTFET_DR_IN) signal fromlow side driver circuit 120. The (BOOTFET_DR_IN) signal may be invertedwith respect to the LS_GATE signal. Bootstrap transistor drive circuit225 may be configured to provide a gate drive signal called (BOOTFET_DR)to a bootstrap transistor in bootstrap charging circuit 157 (see FIG.1), discussed in more detail below. The (BOOTFET_DR) gate drive signalmay be timed to turn on the bootstrap transistor when low sidetransistor 115 is turned on. Also, since bootstrap transistor drivecircuit 225 is driven by (Vcc), the output of this circuit may have avoltage that goes from 0 volts in a low state to (Vcc)+6 volts in a highstate. In one embodiment the bootstrap transistor is turned on after lowside transistor 115 is turned on, and the bootstrap transistor is turnedoff before the low side transistor is turned off.

In some embodiments, the turn on transient of the (BOOTFET_DR) signalmay be delayed by the introduction of a series delay resistor 705 to theinput of second buffer 745, that may be a gate of a transistor in afinal buffer stage. In further embodiments, the turn off transient oflow side transistor 115 (see FIG. 1) may be delayed by the addition of aseries resistor to a gate of a final pull down transistor in low sidedrive circuit 120. In one embodiment, one or more capacitors may be usedin bootstrap transistor drive circuit 225, and support voltages of theorder of (Vcc) which, for example, could be 20 volts, depending on theend user requirements and the design of the circuit. In some embodimentsthe one or more capacitors may be made with a field dielectric to GaNcapacitor instead of a drain to source shorted enhancement-modetransistor.

Now referring to FIG. 8 a block diagram for low side transistor drivecircuit 120 is illustrated. Low side transistor drive circuit 120 mayhave a first inverter 805, a buffer 810, a second inverter 815, a secondbuffer 820 and a third buffer 825. Third buffer 825 may provide the(LS_GATE) signal to low side transistor 115 (see FIG. 1). In someembodiments two inverter/buffer stages may be used because the input tothe gate of low side transistor 115 (see FIG. 1) may be synchronous with(Vin). Thus, (Vin) in a high state may correspond to (Vgate) of low sidetransistor 115 in a high state and vice versa.

In further embodiments, certain portions of low side drive circuit 120may have an asymmetric hysteresis. Some embodiments may includeasymmetric hysteresis using a resistor divider 840 with a transistorpull down 850.

Further embodiments may have multiple input NAND gates for the (STP_LS)signal (shoot through protection on low side transistor 115). In oneembodiment, low side drive circuit 120 may receive the shoot throughprotection signal (STP_LS) from level shift driver circuit 217. Thepurpose of the (STP_LS) signal may be similar to the (STP_HS) signaldescribed previously. The (STP_LS) signal may ensure that low sidetransistor drive circuit 120 does not communicate with gate 117 (seeFIG. 1) of low side transistor 115 when level shift driver circuit 217output is at a high state. In other embodiments, the output of the firstinverter stage 805 may be used as the (STP_HS) signal for level shiftdrive circuit 217 and the (BOOTFET_DR_IN) signal for bootstraptransistor drive circuit 225.

In some embodiments, low side transistor drive circuit 120 may employmultiple input NAND gates for the (LS_UVLO) signal received from UVLOcircuit 227 (see FIG. 2). Further embodiments may employ a turn offdelay resistor that may be in series with a gate of a final pull downtransistor in final buffer stage 825. The delay resistor may be used insome embodiments to make sure the bootstrap transistor is turned offbefore low side transistor 115 turns off.

Now referring to FIG. 9, startup circuit 155 is illustrated in greaterdetail. Startup circuit 155 may be designed to have a multitude offunctionalities as discussed in more detail below. Primarily, startupcircuit 155 may be used to provide an internal voltage (in this caseSTART_Vcc) and provide enough current to support the circuits that arebeing driven by (Vcc). This voltage may remain on to support thecircuits until (Vcc) is charged up to the required voltage externallyfrom rail voltage 135 (V+). Startup circuit 155 may also provide areference voltage (Vref) that may be independent of the startup voltage,and a reference current sink (Iref).

In one embodiment, a depletion-mode transistor 905 may act as theprimary current source in the circuit. In further embodimentsdepletion-mode transistor 905 may be formed by a metal layer disposedover a passivation layer. In some embodiments, depletion-mode transistor905 may use a high voltage field plate (typically intrinsic to anyhigh-voltage GaN technology) as the gate metal. In further embodiments afield dielectric may act as the gate insulator. The resultant gatedtransistor may be a depletion-mode device with a high channel pinch-offvoltage (Vpinch) (i.e., pinch-off voltage is proportional to the fielddielectric thickness). Depletion-mode transistor 905 may be designed toblock relatively high voltages between its drain (connected to V+) andits source. Such a connection may be known as a source followerconnection. Depletion-mode transistor 905 may have a gate 906 coupled toground, a source 907 coupled to a first node 911 and a drain 909 coupledto voltage source 135.

In further embodiments a series of identical diode connectedenhancement-mode low-voltage transistors 910 may be in series withdepletion-mode transistor 905. Series of identical diode connectedenhancement-mode low-voltage transistors 910 may be connected in seriesbetween a first node 911 and a second node 912. One or more intermediatenodes 913 may be disposed between each of series of identical diodeconnected enhancement-mode low-voltage transistors 910. The width tolength ratio of the transistors may set the current drawn from (V+) aswell as the voltage across each diode. To remove threshold voltage andprocess variation sensitivity, series of identical diode connectedenhancement-mode low-voltage transistors 910 may be designed as largechannel length devices. In some embodiments, series of identical diodeconnected enhancement-mode low-voltage transistors 910 may be replacedwith one or more high value resistors.

In further embodiments, at the bottom end of series of identical diodeconnected enhancement-mode low-voltage transistors 910, a current mirror915 may be constructed from two enhancement-mode low-voltage transistorsand used to generate a reference current sink (Iref). First currentmirror transistor 920 may be diode connected and second current mirrortransistor 925 may have a gate connected to the gate of the firstcurrent mirror transistor. The sources of first and second currentmirror transistors 920, 925, respectively may be coupled and tied toground. A drain terminal of first current mirror transistor 920 may becoupled to second junction 912 and a source terminal of second currentmirror transistor 925 may be used as a current sink terminal. This stackof current mirror 915 and series of identical diode connectedenhancement-mode low-voltage transistors 910 may form what is known as a“source follower load” to depletion-mode transistor 905.

In other embodiments, when gate 906 of depletion-mode transistor 905 istied to ground, source 907 of the depletion-mode transistor may assume avoltage close to (Vpinch) when current is supplied to the “sourcefollower load”. At the same time the voltage drop across diode connectedtransistor 920 in current mirror 915 may be close to the thresholdvoltage of the transistor (Vth). This condition implies that the voltagedrop across each of series of identical diode connected enhancement-modelow-voltage transistors 910 may be equal to (Vpinch−Vth)/n where ‘n’ isthe number of diode connected enhancement-mode transistors betweencurrent mirror 915 and depletion-mode transistor 905.

For example, if the gate of a startup transistor 930 is connected to thethird identical diode connected enhancement-mode low-voltage transistorfrom the bottom, the gate voltage of the startup transistor may be3*(Vpinch−Vth)/n+Vth. Therefore, the startup voltage may be3*(Vpinch−Vth)/n+Vth−Vth=3*(Vpinch−Vth)/n. As a more specific example,in one embodiment where (Vpinch)=40 volts, (Vth)=2 volts where n=6 and(Vstartup)=19 volts.

In other embodiments, startup circuit 155 may generate a referencevoltage signal (Vref). In one embodiment, the circuit that generates(Vref) may be similar to the startup voltage generation circuitdiscussed above. A reference voltage transistor 955 may be connectedbetween two transistors in series of identical diode connectedenhancement-mode low-voltage transistors 910. In one embodiment(Vref)=(Vpinch−Vth)/n.

In further embodiments, a disable pull down transistor 935 may beconnected across the gate to source of startup transistor 930. When thedisable signal is high, startup transistor 930 will be disabled. A pulldown resistor 940 may be connected to the gate of disable transistor 935to prevent false turn on of the disable transistor. In other embodimentsa diode clamp 945 may be connected between the gate and the sourceterminals of startup transistor 930 to ensure that the gate to sourcevoltage capabilities of the startup transistor are not violated duringcircuit operation (i.e., configured as gate overvoltage protectiondevices). In some embodiments, diode clamp 945 may be made with a seriesof diode connected GaN-based enhancement-mode transistors 1050, asillustrated in FIG. 10.

Now referring to FIG. 11, UVLO circuit 227 is illustrated in greaterdetail. In some embodiments, UVLO circuit 227 may have a differentialcomparator 1105, a down level shifter 1110 and an inverter 1115. Infurther embodiments, UVLO circuit 227 may use (Vref) and (Iref)generated by startup circuit 155 (see FIG. 9) in a differentialcomparator/down level shifter circuit to generate the (LS_UVLO) signalthat feeds into level shift driver circuit 217 (see FIG. 2) and low sidetransistor driver circuit 120. In some embodiments UVLO circuit 227 canalso be designed to have asymmetric hysteresis. In further embodimentsthe output of UVLO circuit 227 may be independent of threshold voltage.This may be accomplished by choosing a differential comparator with arelatively high gain. In one embodiment the gain can be increased byincreasing the value of the current source and the pull up resistors inthe differential comparator. In some embodiments the limit on thecurrent and resistor may be set by (Vref).

In other embodiments voltages (VA) and (VB), 1120 and 1125,respectively, may be proportional to (Vcc) or (Vdd_LS) and (Vref) asdictated by the resistor divider ratio on each input. When (VA)1120>(VB) 1125 the output of the inverting terminal goes to a low state.In one specific embodiment, the low state=(Vth) since the current sourcecreates a source follower configuration. Similarly when (VA) 1120<(VB)1125 the output goes to a high state (Vref). In some embodiments downlevel shifter 1110 may be needed because the low voltage needs to beshifted down by one threshold voltage to ensure that the low input tothe next stage is below (Vth). The down shifted output may be invertedby a simple resistor pull up inverter 1115. The output of inverter 1115is the (LS_UVLO) signal.

Now referring to FIG. 12, bootstrap capacitor charging circuit 157 isillustrated in greater detail. In one embodiment, bootstrap diode andtransistor circuit 157 may include a parallel connection of a highvoltage diode connected enhancement-mode transistor 1205 and a highvoltage bootstrap transistor 1210. In further embodiments, high voltagediode connected enhancement-mode transistor 1205 and high voltagebootstrap transistor 1210 can be designed to share the same drainfinger. In some embodiments the (BOOTFET_DR) signal may be derived frombootstrap transistor drive circuit 225 (see FIG. 2). As discussed above,high voltage bootstrap transistor 1210 may be turned on coincident withthe turn on of low side transistor 115 (see FIG. 1).

Now referring to FIG. 13, an alternative bootstrap diode and transistorcircuit 1300 may be used in place of bootstrap diode and transistorcircuit 157 discussed above in FIG. 12. In the embodiment illustrated inFIG. 13, a depletion-mode device 1305 cascoded by an enhancement-modelow voltage GaN device 1310 may be connected as illustrated in schematic1300. In another embodiment, a gate of depletion-mode device 1305 can beconnected to ground to reduce the voltage stress on cascodedenhancement-mode device 1310, depending upon the pinch-off voltage ofthe depletion-mode device.

High Side Device

Now referring to FIG. 14, high side logic and control circuit 153 isillustrated in greater detail. In one embodiment, high side driver 130receives inputs from first level shift receiver 1410 and high side UVLOcircuit 1415 and sends a (HS_GATE) signal to high side transistor 125(see FIG. 1). In yet further embodiments, a pull up trigger circuit 1425is configured to receive the (LSHIFT_(—)1) signal and control pull uptransistor 1435. In some embodiments, second level shift receivercircuit 1420 is configured to control blanking transistor 1440. Both thepull up transistor 1435 and blanking transistor 1440 may be connected inparallel with pull up resistor 1430. Each circuit within high side logicand control circuit 153 is discussed below, and in some cases is shownin more detail in FIGS. 16-20.

Now referring to FIG. 15, first level shift receiver 1410 is illustratedin greater detail. In some embodiments, first level shift receiver 1410may convert the (L_SHIFT1) signal to an (LS_HSG) signal that can beprocessed by high side transistor driver 130 (see FIG. 14) to drive highside transistor 125 (see FIG. 1). In further embodiments, first levelshift receiver 1410 may have three enhancement-mode transistors 1505,1510, 1515 employed in a multiple level down shifter and a plurality ofdiode connected transistors 1520 acting as a diode clamp, as discussedin more detail below.

In one embodiment, first level shift receiver 1410 may down shift the(L_SHIFT1) signal by 3*Vth (e.g., each enhancement-mode transistor 1505,1510, 1515 may have a gate to source voltage close to Vth). In someembodiments the last source follower transistor (e.g., in this casetransistor 1515) may have a three diode connected transistor clamp 1520across its gate to source. In further embodiments this arrangement maybe used because its source voltage can only be as high as (Vdd_HS)(i.e., because its drain is connected to Vdd_HS) while its gate voltagecan be as high as V (L_SHIFT1)−2*Vth. Thus, in some embodiments themaximum gate to source voltage on last source follower transistor 1515may be greater than the maximum rated gate to source voltage of thedevice technology. The output of final source follower transistor 1515is the input to high side transistor drive 130 (see FIG. 1), (i.e., theoutput is the LS_HSG signal). In further embodiments fewer or more thanthree source follower transistors may be used. In yet furtherembodiments, fewer or more than three diode connected transistors may beused in clamp 1520.

Now referring to FIG. 16, second level shift receiver 1420 isillustrated in greater detail. In one embodiment, second level shiftreceiver 1420 may have a down level shift circuit 1605 and an invertercircuit 1610. In some embodiments second level shift receiver 1420 maybe constructed in a similar manner as first level shift receiver 1410(see FIG. 15), except the second level shift receiver may have only onedown level shifting circuit (e.g., enhancement-mode transistor 1615) anda follow on inverter circuit 1610. In one embodiment, down level shiftcircuit 1605 may receive the (L_SHIFT2) signal from second level shifttransistor 205 (see FIG. 2). In one embodiment, inverter circuit 1610may be driven by the (Vboot) signal, and the gate voltage of the pull uptransistor of the inverter may be used as the (BLANK_FET) signal drivingblanking transistor 1440 (see FIG. 14). In some embodiments the voltagemay go from 0 volts in a low state to (Vboot+0.5*(Vboot−Vth)) in a highstate. Similar to first level shift receiver 1410, second level shiftreceiver 1420 may have a diode connected transistor clamp 1620 acrossthe gate to source of source follower transistor 1615. In otherembodiments, clamp 1620 may include fewer or more than three diodeconnected transistors.

Now referring to FIG. 17, pull up trigger circuit 1425 is illustrated ingreater detail. In one embodiment, pull up trigger circuit 1425 may havea first inverter 1705, a second inverter 1710, an RC pulse generator1715 and a gate to source clamp 1720. In some embodiments pull uptrigger circuit 1425 may receive the (L_SHIFT1) signal as an input, andin response, generate a pulse as soon as the (L_SHIFT1) voltagetransitions to approximately the input threshold of first inverter 1705.The generated pulse may be used as the (PULLUP_FET) signal that drivespull up transistor 1435 (see FIG. 14). Second inverter 1710 may bedriven by (Vboot) instead of (Vdd_HS) because pull up transistor 1435gate voltage may need to be larger than the (L_SHIFT1) signal voltage.

Now referring to FIG. 18, high side UVLO circuit 1415 is illustrated ingreater detail. In one embodiment, high side UVLO circuit 1415 may havedown level shifter 1805, a resistor pull up inverter with asymmetrichysteresis 1810 and a gate to source clamp 1815. In further embodiments,the (HS_UVLO) signal generated by high side UVLO circuit 1415 may aid inpreventing circuit failure by turning off the (HS_GATE) signal generatedby high side drive circuit 130 (see FIG. 14) when bootstrap capacitor110 voltage goes below a certain threshold. In some embodiments,bootstrap capacitor 110 voltage (Vboot) (i.e., a floating power supplyvoltage) is measured, and in response, a logic signal is generated andcombined with the output signal (LS_HSG) from first level shift receiver1410 which is then used as the input to the high side gate drive circuit130. More specifically, in this embodiment, for example, the UVLOcircuit is designed to engage when (Vboot) reduces to less than 4*Vthabove switch node (Vsw) 145 voltage. In other embodiments a differentthreshold level may be used.

In further embodiments, high side UVLO circuit 1415 may down shift(Vboot) in down level shifter 1805 and transfer the signal to inverterwith asymmetric hysteresis 1810. The output of inverter with asymmetrichysteresis 1810 may generate the (HS_UVLO) signal which is logicallycombined with the output from the first level shift receiver 1410 toturn off high side transistor 125 (see FIG. 1). In some embodiments thehysteresis may be used to reduce the number of self-triggered turn onand turn off events of high side transistor 125 (see FIG. 1), that maybe detrimental to the overall performance of half bridge circuit 100.

Now referring to FIG. 19, high side transistor driver 130 is illustratedin greater detail. High side transistor driver 130 may have a firstinverter stage 1905 followed by a high side drive stage 1910. Firstinverter stage 1905 may invert the down shifted (LS_HSG) signal receivedfrom level shift 1 receiver 1410 (see FIG. 15). The downshifted signalmay then be sent through high side drive stage 1910. High side drivestage 1910 may generate the (HS_GATE) signal to drive high sidetransistor 125 (see FIG. 1). In further embodiments first inverter stage1905 may contain a two input NOR gate that may ensure high sidetransistor 125 (see FIG. 1) is turned off when the (HS_UVLO) signal isin a high state.

Now referring to FIG. 20, a reference voltage generation circuit 2000may be used, to generate a high side reference voltage from a supplyrail. Such a circuit maybe placed on the high side GaN device 105 forgenerating internal power supplies which are referenced to the switchnode voltage 145. In some embodiments, circuit 2000 may be similar tostartup circuit 155 in FIG. 9. One difference in circuit 2000 may be theaddition of a source follower capacitor 2010 connected between firstnode 2011 and second node 2012. In some embodiments, source followercapacitor 2010 may be needed to ensure that a well regulated voltage,which does not fluctuate with dv/dt appearing at the switch node (Vsw)145, develops between the first node 2011 and the second node 2012. Inother embodiments a reference voltage capacitor 2015 may be connectedbetween a source of reference voltage transistor 2055 and second node2012. In some embodiments the drain of the reference voltage transistor2055 may be connected to the (Vboot) node. In some embodiments,reference voltage capacitor 2015 may be needed to ensure that (Vref) iswell regulated and does not respond to high dv/dt conditions at switchnode (Vsw) 145 (see FIG. 1). In yet further embodiments, anotherdifference in circuit 2000 may be that second node 2012 may be coupledto a constantly varying voltage, such as switch node (Vsw) 145 (see FIG.1), rather than a ground connection through a current sink circuit 915(see FIG. 9). In yet further embodiments (Vref) can be used as (Vdd_HS)in the half bridge circuit 100.

Another difference in circuit 2000 may be the addition of a high-voltagediode connected transistor 2025 (i.e., the gate of the transistor iscoupled to the source of the transistor) coupled between depletion-modetransistor 2005 and series of identical diode connected enhancement-modelow-voltage transistors 2020. More specifically, high-voltage diodeconnected transistor 2025 may have source coupled to the source ofdepletion-mode transistor 2005, a drain coupled to first node 2011 and agate coupled to its source. High-voltage diode connected transistor 2025may be used to ensure that source follower capacitor 2010 does notdischarge when the voltage at the top plate of the source followercapacitor rises above (V+). In further embodiments source followercapacitor 2010 may be relatively small and may be integrated on asemiconductor substrate or within an electronic package. Also shown inFIG. 21 is bootstrap capacitor 110 that may be added externally in ahalf bridge circuit.

In some embodiments, shield capacitor 160 (see FIG. 1) may be connectedfrom first level shift node 305 (see FIG. 3) and second level shift node(not shown) to switch node 145 to assist in reducing the falsetriggering discussed above. In some embodiments, the larger the value ofshield capacitor 160, the more immune the circuit will be to falsetriggering effects due to the parasitic capacitance to ground. However,during high side transistor 125 turn off, shield capacitor 160 may bedischarged through pull up resistor 303 (see FIG. 3) connected to firstlevel shift node 305. This may significantly slow down high sidetransistor 125 turn off process. In some embodiments this considerationmay be used to set an upper limit on the value of shield capacitor 160.In further embodiments, an overvoltage condition on first level shiftnode 305 (see FIG. 3) may be prevented by the use of a clamp circuit 161(see FIG. 1) between the first level shift node and switch node 145. Insome embodiments, clamp circuit 161 maybe composed of a diode connectedtransistor where a drain of the transistor is connected to first levelshift node 305 (see FIG. 3) and a gate and a source are connected toswitch node (Vsw) 145 (see FIG. 1). In further embodiments, a secondshield capacitor and a second clamp circuit may be placed between thesecond level shift node and switch node (Vsw) 145 (see FIG. 1).

Half Bridge Circuit #1 Operation

The following operation sequence for half-bridge circuit 100 is forexample only and other sequences may be used without departing from theinvention. Reference will now be made simultaneously to FIGS. 1, 2 and14.

In one embodiment, when the (PWM_LS) signal from the controller is high,low side logic, control and level shift circuit 150 sends a high signalto low side transistor driver 120. Low side transistor driver 120 thencommunicates through the (LS_GATE) signal to low side transistor 115 toturn it on. This will set the switch node voltage (Vsw) 145 close to 0volts. When low side transistor 115 turns on, it provides a path forbootstrap capacitor 110 to become charged through bootstrap chargingcircuit 157 which may be connected between (Vcc) and (Vboot). Thecharging path has a parallel combination of a high voltage bootstrapdiode 1205 (see FIG. 12) and transistor 1210. The (BOOTFET_DR) signalprovides a drive signal to bootstrap transistor 1210 (see FIG. 12) thatprovides a low resistance path for charging bootstrap capacitor 110.

Bootstrap diode 1205 (see FIG. 12) may be used to ensure that there is apath for charging bootstrap capacitor 110 during startup when there isno low side transistor 115 gate drive signal (LS_GATE). During this timethe (PWM_HS) signal should be low. If the (PWM_HS) signal isinadvertently turned on (i.e., in a high state) during this time the(STP_HS) signal generated from low side transistor driver 120 willprevent high side transistor 125 from turning on. If the (PWM_LS) signalis turned on while the (PWM_HS) signal is on, the (STP_LS) signalgenerated from level shift driver circuit 217 will prevent low sidetransistor 115 from turning on. Also, in some embodiments the (LS_UVLO)signal may prevent low side transistor 115 and high side transistor 125from turning on when either (Vcc) or (Vdd_LS) goes below a presetthreshold voltage level.

In further embodiments, when the (PWM_LS) signal is low, low side gatesignal (LS_GATE) to low side transistor 115 is also low. During the deadtime between the (PWM_LS) signal low state to the (PWM_HS) high statetransition, an inductive load will force either high side transistor 125or low side transistor 115 to turn on in the synchronous rectifier mode,depending on direction of power flow. If high side transistor 125 turnson during the dead time (e.g., during boost mode operation), switch node(Vsw) 145 voltage may rise close to (V+) 135 (rail voltage).

In some embodiments, a dv/dt condition on switch node 145 (Vsw) may tendto pull first level shift node (LSHIFT_(—)1) 305 (see FIG. 3) to a lowstate relative to switch node (Vsw) 145, due to capacitive coupling toground. This may turn on high side gate drive circuit 130 causingunintended triggering of high side transistor 125. In one embodiment,this may result in no dead time which may harm half bridge circuit 100with a shoot through condition. In further embodiments, to prevent thiscondition from occurring, blanking pulse generator 223 may sense theturn off transient of low side transistor 115 and send a pulse to turnon second level shift transistor 205. This may pull the (L_SHIFT2)signal voltage to a low state which then communicates with second levelshift receiver 1420 to generate a blanking pulse signal (B_PULSE) todrive blanking transistor 1440. Blanking transistor 1440 may then act asa pull up to prevent first level shift node (LSHIFT_(—)1) 305 (see FIG.3) from going to a low state relative to switch node (Vsw) 145.

In further embodiments, after the dead time, when the (PWM_HS) signalgoes to a high state, level shift driver circuit 217 may send a highsignal to the gate of first level shift transistor 203 (via the L1_DRsignal from level shift driver circuit 217). The high signal will pullfirst level shift node (LSHIFT_(—)1) 305 (see FIG. 3) low relative toswitch node (Vsw) 145 which will result in a high signal at the input ofhigh side transistor 125, turning on high side transistor 125. Switchnode voltage (Vsw) 145 will remain close to (V+) 135. In one embodiment,during this time, bootstrap capacitor 110 may discharge through firstlevel shift transistor 203 (which is in an on state during this time).

If high side transistor 125 stays on for a relatively long time (i.e., alarge duty cycle) bootstrap capacitor 110 voltage will go down to a lowenough voltage that it will prevent high side transistor 125 fromturning off when the (PWM_HS) signal goes low. In some embodiments thismay occur because the maximum voltage the (L_SHIFT1) signal can reach is(Vboot) which may be too low to turn off high side transistor 125. Insome embodiments, this situation may be prevented by high side UVLOcircuit 1415 that forcibly turns off high side transistor 125 by sendinga high input to high side gate drive circuit 130 when (Vboot) goes belowa certain level.

In yet further embodiments, when the (PWM_HS) signal goes low, firstlevel shift transistor 203 will also turn off (via the L1_DR signal fromthe level shift driver circuit 217). This will pull first level shiftnode (LSHIFT_(—)1) 305 (see FIG. 3) to a high state. However, in someembodiments this process may be relatively slow because the high valuepull up resistor 303 (see FIG. 3) (used to reduce power consumption insome embodiments) needs to charge all the capacitances attached to firstlevel shift node (L_SHIFT1) 305 (see FIG. 3) including the outputcapacitance (Coss) of first level shift transistor 213 and shieldcapacitor 160. This may increase the turn off delay of high sidetransistor 125. In order to reduce high side transistor 125 turn offdelay, pull up trigger circuit 1425 may be used to sense when firstlevel shift node (L_SHIFT1) 305 (see FIG. 3) goes above (Vth). Thiscondition may generate a (PULLUP_FET) signal that is applied to pull uptransistor 1435 which, acting in parallel with pull up resistor 1430,may considerably speed up the pull up of first level shift node(L_SHIFT1) 305 (see FIG. 3) voltage, hastening the turn off process.

Half Bridge Circuit #2

Now referring to FIG. 21, a second embodiment of a half bridge circuit2100 is disclosed. Half bridge circuit 2100 may have the same blockdiagram as circuit 100 illustrated in FIG. 1, however the level shifttransistors in circuit 2100 may operate with pulsed inputs, rather thana continuous signal, as described in more detail below. In someembodiments, pulsed inputs may result in lower power dissipation,reduced stress on the level shift transistors and reduced switchingtime, as discussed in more detail below.

Continuing to refer to FIG. 21, one embodiment includes an integratedhalf bridge power conversion circuit 2100 employing a low side GaNdevice 2103, a high side GaN device 2105, a load 2107, a bootstrapcapacitor 2110 and other circuit elements, as discussed in more detailbelow. Some embodiments may also have an external controller (not shownin FIG. 21) providing one or more inputs to circuit 2100 to regulate theoperation of the circuit. Circuit 2100 is for illustrative purposes onlyand other variants and configurations are within the scope of thisdisclosure.

As further illustrated in FIG. 21, in one embodiment, integrated halfbridge power conversion circuit 2100 may include a low side circuitdisposed on low side GaN device 2103 that includes a low side transistor2115 having a low side control gate 2117. The low side circuit mayfurther include an integrated low side transistor driver 2120 having anoutput 2123 connected to a low side transistor control gate 2117. Inanother embodiment there may be a high side circuit disposed on highside GaN device 2105 that includes a high side transistor 2125 having ahigh side control gate 2127. The high side circuit may further includean integrated high side transistor driver 2130 having an output 2133connected to high side transistor control gate 2127.

High side transistor 2125 may be used to control the power input intopower conversion circuit 2100 and have a voltage source (V+) 2135(sometimes called a rail voltage) connected to a drain 2137 of the highside transistor. High side transistor 2125 may further have a source2140 that is coupled to a drain 2143 of low side transistor 2115,forming a switch node (Vsw) 2145. Low side transistor 2115 may have asource 2147 connected to ground. In one embodiment, low side transistor2115 and high side transistor 2125 may be enhancement-mode field-effecttransistors. In other embodiments low side transistor 2115 and high sidetransistor 2125 may be any other type of device including, but notlimited to, GaN-based depletion-mode transistors, GaN-baseddepletion-mode transistors connected in series with silicon basedenhancement-mode field-effect transistors having the gate of thedepletion-mode transistor connected to the source of the silicon-basedenhancement-mode transistor, silicon carbide based transistors orsilicon-based transistors.

In some embodiments high side device 2105 and low side device 2103 maybe made from a GaN-based material. In one embodiment the GaN-basedmaterial may include a layer of GaN on a layer of silicon. In furtherembodiments the GaN based material may include, but not limited to, alayer of GaN on a layer of silicon carbide, sapphire or aluminumnitride. In one embodiment the GaN based layer may include, but notlimited to, a composite stack of other III nitrides such as aluminumnitride and indium nitride and III nitride alloys such as AlGaN andInGaN

Low Side Device

Low side device 2103 may have numerous circuits used for the control andoperation of the low side device and high side device 2105. In someembodiments, low side device 2103 may include a low side logic, controland level shift circuit (low side control circuit) 2150 that controlsthe switching of low side transistor 2115 and high side transistor 2125along with other functions, as discussed in more detail below. Low sidedevice 2103 may also include a startup circuit 2155, a bootstrapcapacitor charging circuit 2157 and a shield capacitor 2160, as alsodiscussed in more detail below.

Now referring to FIG. 22, the circuits within low side control circuit2150 are functionally illustrated. Each circuit within low side controlcircuit 2150 is discussed below, and in some cases is shown in moredetail in FIGS. 23-28. In one embodiment the primary function of lowside control circuit 2150 may be to receive one or more input signals,such as a PWM signal from a controller, and control the operation of lowside transistor 2115, and high side transistor 2125.

First level shift transistor 2203, may be an “on” pulse level shifttransistor, while second level shift transistor 2215 may be an “off”pulse level shift transistor. In one embodiment, a pulse width modulatedhigh side (PWM_HS) signal from a controller (not shown) may be processedby inverter/buffer 2250 and sent on to an on pulse generator 2260 and anoff pulse generator 2270. On pulse generator 2260 may generate a pulsethat corresponds to a low state to high state transient of the (PWM_HS)signal, thus turning on first level shift transistor 2203 during theduration of the pulse. Off pulse generator 2270 may similarly generate apulse that corresponds to the high state to low state transition of the(PWM_HS) signal, thus turning on second level shift transistor 2205 forthe duration of the off pulse.

First and second level shift transistors 2203, 2205, respectively, mayoperate as pull down transistors in resistor pull up inverter circuits.More specifically, turning on may mean the respective level shift nodevoltages get pulled low relative to switch node (Vsw) 2145 voltage, andturning off may result in the respective level shift nodes assuming the(Vboot) voltage. Since first and second level shift transistors 2203,2215, respectively, are “on” only for the duration of the pulse, thepower dissipation and stress level on these two devices may be less thanhalf bridge circuit 100 illustrated in FIG. 1.

First and second resistors 2207, 2208, respectively, may be added inseries with the sources of first and second level shift transistors2203, 2215, respectively to limit the gate to source voltage andconsequently the maximum current through the transistors. First andsecond resistors 2207, 2208, respectively, could be smaller than thesource follower resistors in half bridge circuit 100 illustrated in FIG.1, which may help make the pull down action of first and second levelshift transistors 2203, 2215 faster, reducing the propagation delays tohigh side transistor 2125.

In further embodiments, first and second resistors 2207, 2208,respectively, could be replaced by any form of a current sink. Oneembodiment may connect the source of first and second level shifttransistors 2203, 2205, respectively to a gate to source shorteddepletion-mode device. One embodiment of a depletion-mode transistorformed in a high-voltage GaN technology may be to replace theenhancement-mode gate stack with one of the high-voltage field platemetals superimposed on top of the field dielectric layers. The thicknessof the field dielectric and the work function of the metal may controlthe pinch-off voltage of the stack.

In further embodiments, first and second resistors 2207, 2208,respectively may be replaced by a current sink. In one embodiment areference current (Iref) that is generated by startup circuit 2155 (seeFIG. 21) may be used. Both the depletion-mode transistor and currentsink embodiments may result in a significant die area reduction comparedto the resistor option (i.e., because a small depletion transistor wouldsuffice and Iref is already available).

Bootstrap transistor drive circuit 2225 may be similar to bootstraptransistor drive circuit 225 illustrated in FIG. 2 above. Bootstraptransistor drive circuit 2225 may receive input from low side drivecircuit 2220 (see FIG. 22) and provide a gate drive signal called(BOOTFET_DR) to the bootstrap transistor in bootstrap capacitor chargingcircuit 2157 (see FIG. 21), as discussed in more detail above.

Now referring to FIG. 23, first level shift transistor 2203 isillustrated along with a pull up resistor 2303 that may be located inhigh side device 2105. In some embodiments, first level shift transistor2203 may operate as a pull down transistor in a resistor pull upinverter similar to first level shift transistor 203 illustrated in FIG.3. As discussed above, pull up resistor 2303 may be disposed in highside device 2105 (see FIG. 21). Second level shift transistor 2215 mayhave a similar configuration. In some embodiments there may be a firstcapacitance between the first output terminal (LS_NODE) 2305 and switchnode (Vsw) 2145 (see FIG. 21), and a second capacitance between a firstoutput terminal 2305 and ground, where the first capacitance is greaterthan the second capacitance. The first capacitance may be designed suchthat in response to a high dv/dt signal at the switch node (Vsw) 2145(see FIG. 21), a large portion of the C*dv/dt current is allowed toconduct through the first capacitance ensuring that the voltage at firstoutput terminal 2305 tracks the voltage at the switch node (Vsw). Ashield capacitor 2160 (see FIG. 21) may be configured to act as thefirst capacitor as described above. In further embodiments shieldcapacitor 2160 (see FIG. 21) may be used to create capacitance betweenfirst output terminal 2305 and switch node (Vsw) 2145 (see FIG. 21) inthe half bridge power conversion circuit 2100. Shield capacitor 2160 mayalso be used to minimize the capacitance between first output terminal2305 and a substrate of the semiconductor device. In further embodimentsshield capacitor 2160 may be constructed on low side GaN device 2103.

Now referring to FIG. 24, inverter/buffer circuit 2250 is illustrated ingreater detail. In one embodiment inverter/buffer circuit 2250 may havea first inverter stage 2405 and a first buffer stage 2410. In furtherembodiments, inverter/buffer circuit 2250 may be driven directly by the(PWM_HS) signal from the controller (not shown). The output of firstinverter stage 2405 may be the input signal (PULSE_ON) to on pulsegenerator 2260 (see FIG. 22) while the output of first buffer stage 2410may be an input signal (PULSE_OFF) to off pulse generator 2270.

In some embodiments, an optional (LS_UVLO) signal may be generated bysending a signal generated by UVLO circuit 2227 (see FIG. 22) in to aNAND gate disposed in first inverter stage 2405. This circuit may beused to turn off the level shift operation if either (Vcc) or (Vdd_LS)go below a certain reference voltage (or a fraction of the referencevoltage). In further embodiments, inverter/buffer circuit 2250 may alsogenerate a shoot through protection signal (STP_LS1) for low sidetransistor 2115 (see FIG. 21) that may be applied to low side transistorgate drive circuit 2120. This may turn off low side transistor gatedrive circuit 2120 (see FIG. 21) when the (PWM_HS) signal is high,preventing shoot through.

Now referring to FIG. 25, on pulse generator 2260 is illustrated ingreater detail. In one embodiment on pulse generator 2260 may have afirst inverter stage 2505, a first buffer stage 2510, an RC pulsegenerator 2515, a second inverter stage 2520 a third inverter stage 2525and a third buffer stage 2530. In further embodiments the (PULSE_ON)signal input from inverter/buffer circuit 2250 (see FIG. 22) may befirst inverted and then transformed into an on pulse by RC pulsegenerator 2515 and a square wave generator. The result of this operationis the gate drive signal (LI_DR) that is transmitted to first levelshift transistor 2203 (see FIG. 22).

In further embodiments, on pulse generator 2260 may comprise one or morelogic functions, such as for example, a binary or combinatorialfunction. In one embodiment, on pulse generator 2260 may have a multipleinput NOR gate for the (STP_HS) signal. The (STP_HS) signal may have thesame polarity as the (LS_GATE) signal. Therefore, if the (STP_HS) signalis high (corresponding to LS_GATE signal being high) the on pulse maynot be generated because first inverter circuit 2505 in FIG. 25 will bepulled low which will deactivate pulse generator 2515.

In further embodiments, RC pulse generator 2515 may include a clampdiode (not shown). The clamp diode may be added to ensure that RC pulsegenerator 2515 works for very small duty cycles for the (PWM_LS) signal.In some embodiments, on pulse generator 2260 may be configured toreceive input pulses in a range of 2 nanoseconds to 20 microseconds andto transmit pulses of substantially constant duration within the range.In one embodiment the clamp diode may turn on and short out a resistorin RC pulse generator 2515 (providing a very small capacitor dischargetime) if the voltage across the clamp diode becomes larger than (Vth).This may significantly improve the maximum duty cycle of operation (withrespect to the PWM_HS signal) of pulse generator circuit 2260.

Now referring to FIG. 26, off pulse generator 2270 is illustrated ingreater detail. In one embodiment off pulse generator 2270 may have anRC pulse generator 2603, a first inverter stage 2605, a second inverterstage 2610 and a first buffer stage 2615. In further embodiments, offpulse generator 2270 may receive an input signal (PULSE_OFF) frominverter/buffer circuit 2250 (see FIG. 22) that may be subsequentlycommunicated to RC pulse generator 2603.

In further embodiments the pulse from RC pulse generator 2603 is sentthrough first inverter stage 2605, second inverter stage 2610 and bufferstage 2615. The pulse may then be sent as the (L2_DR) signal to secondlevel shift transistor 2215 (see FIG. 22). A clamp diode may also beincluded in off pulse generator 2270. In some embodiments, the operatingprinciple may be similar to the operating principle discussed above withregard to on pulse generator 2260 (see FIG. 25). Such operatingprinciples may ensure that off pulse generator 2270 operates for verylow on times of high side transistor 2125 (see FIG. 21) (i.e. thecircuit will operate for relatively small duty cycles). In someembodiments, off pulse generator 2270 may be configured to receive inputpulses in a range of 2 nanoseconds to 20 microseconds and to transmitpulses of substantially constant duration within the range. In furtherembodiments an off level shift pulse can be shortened by an on inputpulse to enable an off time of less than 50 nanoseconds on high sidetransistor 2125.

In some embodiments, RC pulse generator 2603 may include a capacitorconnected with a resistor divider network. The output from the resistormay be a signal (INV) that is sent to an inverter 2275 (see FIG. 22)that generates a shoot through protection signal (STP_LS2) transmittedto low side driver circuit 2220. In further embodiments, off pulsegenerator 2270 may comprise one or more logic functions, such as forexample, a binary or combinatorial function. In one embodiment the(STP_LS2) signal is sent to a NAND logic circuit within low side drivercircuit 2220, similar to the (STP_LS1) signal. In some embodiments,these signals may be used to ensure that during the duration of the offpulse signal (PULSE_OFF), low side transistor 2115 (see FIG. 21) doesnot turn on (i.e., because high side transistor 2125 turns off duringthe off pulse). In some embodiments this methodology may be useful tocompensate for a turn off propagation delay (i.e., the PULSE_OFF signalmay enable shoot through protection), ensuring that low side transistor2115 will only turn on after high side transistor 2125 gate completelyturns off.

In further embodiments, a blanking pulse can be level shifted to highside device 2105 using second level shift transistor 2215. To accomplishthis, a blanking pulse may be sent into a NOR input into first inverterstage 2605. The blanking pulse may be used to inhibit false triggeringdue to high dv/dt conditions at switch node Vsw 2145 (see FIG. 20). Insome embodiments no blanking pulse may be used to filter dv/dt inducedor other unwanted level shift output pulses.

Now referring to FIG. 27, blanking pulse generator 2223 is illustratedin greater detail. In one embodiment, blanking pulse generator 2223 maybe a more simple design than used in half bridge circuit 100 illustratedin FIG. 1 because the square wave pulse generator is already part of offpulse generator 2270. In one embodiment the (LS_GATE) signal is fed asthe input to blanking pulse generator 2223 from low side gate drivecircuit 2220 (see FIG. 22). This signal may be inverted and then sentthrough an RC pulse generator to generate a positive going pulse. Insome embodiments, an inverted signal may be used because the pulse needsto correspond to the falling edge of the (LS_GATE) signal. The output ofthis may be used as the blanking pulse input (B_PULSE) to off pulsegenerator 2270.

Now referring to FIG. 28, low side transistor drive circuit 2220 isillustrated in greater detail. In one embodiment low side transistordrive circuit 2220 may have a first inverter stage 2805, a first bufferstage 2810, a second inverter stage 2815, a second buffer stage 2820 anda third buffer stage 2825. In some embodiments two inverter/bufferstages may be used because the input to the gate of low side transistor2115 is synchronous with the (PWM_LS) signal. Thus, in some embodimentsa (PWM_LS) high state may correspond to a (LS_GATE) high state and viceversa.

In further embodiments, low side transistor drive circuit 2220 may alsoinclude an asymmetric hysteresis using a resistor divider with atransistor pull down similar to the scheme described in 120 (see FIG.8). In one embodiment low side transistor drive circuit 2220 includesmultiple input NAND gates for the (STP_LS1) and (STP_LS2) (shoot throughprevention on low side transistor 2115) signals. The (STP_LS1) and(STP_LS2) signals may ensure that low side transistor drive circuit 2220(see FIG. 22) does not communicate with low side transistor 2115 (seeFIG. 21) when high side transistor 2125 is on. This technique may beused to avoid the possibility of shoot-through. Other embodiments mayinclude NAND gates (similar to the ones employed above in FIG. 28) forthe (LS_UVLO) signal. One embodiment may include a turn off delayresistor in series with the gate of the final pull down transistor. Thismay be used to ensure the bootstrap transistor is turned off before lowside transistor 2115 turns off.

In further embodiments, low side device 2103 (see FIG. 21) may alsoinclude a startup circuit 2155, bootstrap capacitor charging circuit2157, a shield capacitor 2160, and a UVLO circuit 2227 that may besimilar to startup circuit 155, bootstrap capacitor charging circuit157, shield capacitor 160 and UVLO circuit 227, respectively, asdiscussed above.

High Side Device

Now referring to FIG. 29, high side logic and control circuit 2153 andhow it interacts with high side transistor driver 2130 is illustrated ingreater detail. In some embodiments, high side logic and control circuit2153 may operate in similar ways as high side logic and control circuit153, discussed above in FIG. 15. In further embodiments, high side logicand control circuit 2153 may operate in different ways, as discussed inmore detail below.

In one embodiment, level shift 1 receiver circuit 2910 receives an(L_SHIFT1) signal from first level shift transistor 2203 (see FIG. 22)that receives an on pulse at the low state to high state transition ofthe (PWM_HS) signal, as discussed above. In response, level shift 1receiver circuit 2910 drives a gate of pull up transistor 2960 (e.g., insome embodiments a low-voltage enhancement-mode GaN transistor). Infurther embodiments, pull up transistor 2960 may then pull up a statestoring capacitor 2955 voltage to a value close to (Vdd_HS) with respectto switch node (Vsw) 2145 voltage. The voltage on a state storingcapacitor 2955 may then be transferred to high side transistor driver2130 and on to the gate of high side transistor gate 2127 (see FIG. 21)to turn on high side transistor 2125. In some embodiments state storingcapacitor 2955 may be a latching storage logic circuit configured tochange state in response to a first pulsed input signal and to changestate in response to a second pulsed input signal. In furtherembodiments, state storing capacitor 2955 may be replaced by any type ofa latching circuit such as, but not limited to an RS flip-flop.

In further embodiments, during this time, level shift 2 receiver circuit2920 may maintain pull down transistor 2965 (e.g., in some embodiments alow-voltage enhancement-mode GaN transistor) in an off state. This maycut off any discharge path for state storing capacitor 2955. Thus, insome embodiments, state storing capacitor 2955 may have a relativelysmall charging time constant and a relatively large discharge timeconstant.

Similarly, level shift 2 receiver 2920 may receive an (L_SHIFT2) signalfrom second level shift transistor 2215 (see FIG. 22) that receives anoff pulse at the high state to low state transition of the (PWM_HS)signal, as discussed above. In response, level shift 2 receiver circuit2920 drives a gate of pull down transistor 2965 (e.g., in someembodiments a low-voltage enhancement-mode GaN transistor). In furtherembodiments, pull down transistor 2965 may then pull down (i.e.,discharge) state storing capacitor 2955 voltage to a value close toswitch node (Vsw) 2145, that may consequently turn off high sidetransistor 2125 through high side transistor driver 2130.

Continuing to refer to FIG. 29, first and second shield capacitors 2970,2975, respectively, may be connected from (L_SHIFT1) and (L_SHIFT2)nodes to help prevent false triggering during high dv/dt conditions atswitch node (Vsw) 2145 (see FIG. 21). In further embodiments there mayalso be a clamp diode between the (L_SHIFT1) and (L_SHIFT2) nodes andthe switch node (Vsw) 2145 (see FIG. 21). This may ensure that thepotential difference between switch node (Vsw) 2145 (see FIG. 21) andthe (L_SHIFT1) and (L_SHIFT2) nodes never goes above (Vth). This may beused to create a relatively fast turn on and turn off for high sidetransistor 2125 (see FIG. 21).

Now referring to FIG. 30, level shift 1 receiver 2910 is illustrated ingreater detail. In one embodiment level shift 1 receiver 2910 mayinclude a down level shifter 3005, a first inverter 3010, a secondinverter 3015, a first buffer 3020, a third inverter 3025, a secondbuffer 3030 and a third buffer 3135. In some embodiments, level shift 1receiver 2910 down shifts (i.e., modulates) the (L_SHIFT1) signal by avoltage of 3*Vth (e.g., using three enhancement-mode transistors whereeach may have a gate to source voltage close to Vth). In otherembodiments a fewer or more downshift transistors may be used.

In further embodiments, the last source follower transistor may have athree diode connected transistor clamp across its gate to its source. Insome embodiments this configuration may be used because its sourcevoltage can only be as high as (Vdd_HS) (i.e., because its drain isconnected to Vdd_HS) while its gate voltage can be as high as V(L_SHIFT1)−2*Vth. Thus, in some embodiments the maximum gate to sourcevoltage on the final source follower transistor can be greater than themaximum rated gate to source voltage in the technology.

In further embodiments, first inverter 3010 may also have a NOR Gate forthe high side under voltage lock out using the (UV_LS1) signal generatedby high side UVLO circuit 2915. In one embodiment, an output of levelshift 1 receiver 2910 (see FIG. 29) may be a (PU_FET) signal that iscommunicated to a gate of pull up transistor 2960 (see FIG. 29). Thissignal may have a voltage that goes from 0 volts in a low state to(Vdd_HS)+(Vdd_HS−Vth) in a high state. This voltage may remain on forthe duration of the on pulse.

Now referring to FIG. 31, level shift 2 receiver 2920 is illustrated ingreater detail. In one embodiment level shift 2 receiver 2920 may besimilar to level shift 1 receiver 2910 discussed above. In furtherembodiments level shift 2 receiver 2920 may include a blanking pulsegenerator 3105, a down level shifter 3110, a first inverter 3115, asecond inverter 3120, a first buffer 3125, an third inverter 3130, asecond buffer 3135 and a third buffer 3140. In one embodiment, blankingpulse generator 3105 may be used in addition to a 3*Vth down levelshifter 3110 and multiple inverter/buffer stages.

In other embodiments different configurations may be used. In someembodiments, this particular configuration may be useful when levelshift 2 receiver 2920 doubles as a high side transistor 2125 (see FIG.21) turn off as well as a blanking transistor 2940 (see FIG. 29) drivefor better dv/dt immunity. In some embodiments, blanking pulse generator3105 may be identical to level shift 2 receiver 1520 illustrated in FIG.17. In one embodiment level shift 2 receiver 2920 (see FIG. 29) mayreceive (L_SHIFT2) and (UV_LS2) signals and in response, transmit a(PD_FET) signal to pull down transistor 2965. In further embodiments,first inverter 3115 may have a two input NAND gate for the (UV_LS2)signal from high side UVLO circuit 2915 (see FIG. 29).

Now referring to FIG. 32, high side UVLO circuit 2915 is illustrated ingreater detail. In one embodiment high side UVLO circuit 2915 mayinclude a down level shifter 3205 and a resistor pull up inverter stage3210. In some embodiments, high side UVLO circuit 2915 may be configuredto prevent circuit failure by turning off the (HS_GATE) signal to highside transistor 2125 (see FIG. 21) when bootstrap capacitor 2110 voltagegoes below a certain threshold. In one example embodiment high side UVLOcircuit 2915 is designed to engage when (Vboot) reduces to a value lessthan 4*Vth below switch node (Vsw) 2145 voltage. In another embodimentthe output of down level shifter 3205 may be a (UV_LS2) signaltransmitted to second level shift receiver 2920 and the output ofresistor pull up inverter stage 3210 may be an (UV_LS1) signal that istransmitted to first level shift receiver 2910.

As discussed below, in some embodiments high side UVLO circuit 2915 maybe different from high side UVLO circuit 1415 for half bridge circuit100 discussed above in FIGS. 14 and 18, respectively. In one embodiment,the (Vboot) signal may be down shifted by 3*Vth and transferred toresistor pull up inverter stage 3210. In further embodiments, sincelevel shift 2 receiver circuit 2920 (see FIG. 29) controls the turn offprocess based on high side transistor 2125 (see FIG. 21), directlyapplying a 3*Vth down shifted output to the NAND gate at the input oflevel shift 2 receiver circuit 2920 will engage the under voltage lockout.

However, in some embodiments, because the bootstrap voltage may be toolow this may also keep pull up transistor 2960 (see FIG. 29) on. In someembodiments, this may result in a conflict. While level shift 2 receivercircuit 2920 (see FIG. 29) tries to keep high side transistor 2125 (seeFIG. 21) off, level shift 1 receiver circuit 2910 may try to turn thehigh side transistor on. In order to avoid this situation, someembodiments may invert the output of the 3*Vth down shifted signal fromhigh side UVLO circuit 2915 (see FIG. 29) and send it to a NOR input onlevel shift 1 receiver circuit 2910. This may ensure that level shift 1receiver circuit 2910 does not interfere with the UVLO induced turn offprocess.

Now referring to FIG. 33, high side transistor driver 2130 isillustrated in greater detail. In one embodiment high side transistordriver 2130 may include a first inverter 3305, a first buffer 3310, asecond inverter 3315, a second buffer 3320 and a third buffer 3325. Insome embodiments high side transistor driver 2130 may be a more basicdesign than high side transistor driver 130 employed in half bridgecircuit 100 illustrated in FIG. 1. In one embodiment, high sidetransistor driver 2130 receives an (S_CAP) signal from state storagecapacitor 2955 (see FIG. 29) and delivers a corresponding drive(HS_GATE) signal to high side transistor 2125 (see FIG. 21). Morespecifically, when the (S_CAP) signal is in a high state, the (HS_GATE)signal is in a high state and vise versa.

Half Bridge Circuit #2 Operation

The following operation sequence for half-bridge circuit 2100 (see FIG.21) is for example only and other sequences may be used withoutdeparting from the invention. Reference will now be made simultaneouslyto FIGS. 21, 22 and 29.

In one embodiment, when the (PWM_LS) signal is in a high state, low sidelogic, control and level shift circuit 2150 may send a high signal tolow side transistor driver 2120 which then communicates that signal tolow side transistor 2115 to turn it on. This may set switch node (Vsw)2145 voltage close to 0 volts. In further embodiments, when low sidetransistor 2115 turns on it may provide a path for bootstrap capacitor2110 to charge. The charging path may have a parallel combination of ahigh-voltage bootstrap diode and transistor.

In some embodiments, bootstrap transistor drive circuit 2225 may providea drive signal (BOOTFET_DR) to the bootstrap transistor that provides alow resistance path for charging bootstrap capacitor 2110. In oneembodiment, the bootstrap diode may ensure that there is a path forcharging bootstrap capacitor 2110 during startup when there is no lowside gate drive signal (LS_GATE). During this time the (PWM_HS) signalshould be in a low state. If the (PWM_HS) signal is inadvertently turnedon during this time, the (STP_HS) signal generated from low side drivercircuit 2220 may prevent high side transistor 2125 from turning on. Ifthe (PWM_LS) signal is turned on while the (PWM_HS) signal is on, thenthe (STP_LS1) and (STP_LS2) signals generated from inverter/buffer 2250and inverter 2275, respectively will prevent low side transistor 2115from turning on. In addition, in some embodiments the (LS_UVLO) signalmay prevent low side gate 2117 and high side gate 2127 from turning onwhen either (Vcc) or (Vdd_LS) go below a predetermined voltage level.

Conversely, in some embodiments when the (PWM_LS) signal is in a lowstate, the (LS_GATE) signal to low side transistor 2115 may also be in alow state. In some embodiments, during the dead time between the(PWM_LS) low signal and the (PWM_HS) high signal transition, theinductive load may force either high side transistor 2125 or low sidetransistor 2115 to turn on in the synchronous rectifier mode, dependingon the direction of power flow. If high side transistor 2125 turns onduring the dead time (e.g., in a boost mode), switch node (Vsw) 2145voltage may rise close to (V+) 2135 (i.e., the rail voltage). This dv/dtcondition on switch node (Vsw) 2145 may tend to pull the (L_SHIFT1) nodeto a low state relative to the switch node (i.e., because of capacitivecoupling to ground) which may turn on high side transistor driver 2130causing unintended conduction of high side transistor 2125. Thiscondition may negate the dead time, causing shoot through.

In some embodiments this condition may be prevented by using blankingpulse generator 2223 to sense the turn off transient of low sidetransistor 2115 and send a pulse to turn on second level shifttransistor 2205. This may pull the (L_SHIFT2) signal to a low statewhich may then communicate with level shift 2 receiver circuit 2920 togenerate a blanking pulse to drive blanking transistor 2940. In oneembodiment, blanking transistor 2940 may act as a pull up to prevent the(L_SHIFT1) signal from going to a low state relative to switch node(Vsw) 2145.

In further embodiments, after the dead time when the (PWM_HS) signaltransitions from a low state to a high state, an on pulse may begenerated by on pulse generator 2260. This may pull the (L_SHIFT1) nodevoltage low for a brief period of time. In further embodiments thissignal may be inverted by level shift 1 receiver circuit 2910 and abrief high signal will be sent to pull up transistor 2960 that willcharge state storage capacitor 2955 to a high state. This may result ina corresponding high signal at the input of high side transistor driver2130 which will turn on high side transistor 2125. Switch node (Vsw)2145 voltage may remain close to (V+) 2135 (i.e., the rail voltage).State storing capacitor 2955 voltage may remain at a high state duringthis time because there is no discharge path.

In yet further embodiments, during the on pulse, bootstrap capacitor2110 may discharge through first level shift transistor 2203. However,since the time period is relatively short, bootstrap capacitor 2110 maynot discharge as much as it would if first level shift transistor 2203was on during the entire duration of the (PWM_HS) signal (as was thecase in half bridge circuit 100 in FIG. 1). More specifically, in someembodiments this may result in the switching frequency at which the UVLOengages to be a relatively lower value than in half bridge circuit 100in FIG. 1.

In some embodiments, when the (PWM_HS) signal transitions from a highstate to a low state, an off pulse may be generated by off pulsegenerator 2270. This may pull the (L_SHIFT2) node voltage low for abrief period of time. This signal may be inverted by level shift 2receiver circuit 2920 and a brief high state signal may be sent to pulldown transistor 2965 that will discharge state storing capacitor 2955 toa low state. This will result in a low signal at the input of high sidetransistor driver 2130 that will turn off high side transistor 2125. Infurther embodiments, state storing capacitor 2955 voltage may remain ata low state during this time because it has no discharge path.

In one embodiment, since the turn off process in circuit 2100 does notinvolve charging level shift node capacitors through a high value pullup resistor, the turn off times may be relatively shorter than in halfbridge circuit 100 in FIG. 1. In further embodiments, high sidetransistor 2125 turn on and turn off processes may be controlled by theturn on of substantially similar level shift transistors 2203, 2205,therefore the turn on and turn off propagation delays may besubstantially similar. This may result in embodiments that have no needfor a pull up trigger circuit and/or a pull up transistor as were bothused in half bridge circuit 100 in FIG. 1.

ESD Circuits

Now referring to FIG. 34, in some embodiments, one or more pins (i.e.,connections from a semiconductor device within an electronic package toan external terminal on the electronic package) may employ anelectro-static discharge (ESD) clamp circuit to protect the circuit. Thefollowing embodiments illustrate ESD clamp circuits that may be used onone or more pins in one or more embodiments disclosed herein, as well asother embodiments that may require ESD protection. In furtherembodiments, the ESD clamp circuits disclosed herein may be employed onGaN-based devices.

One embodiment of an electro-static discharge (ESD) clamp circuit 3400is illustrated. ESD clamp circuit 3400 may have a configurationemploying one or more source follower stages 3405 made fromenhancement-mode transistors. Each source follower stage 3405 may have agate 3406 connected to a source 3407 of an adjacent source followerstage. In the embodiment illustrated in FIG. 34, four source followerstages 3405 are employed, however in other embodiments fewer or more maybe used. Resistors 3410 are coupled to sources 3407 of source followerstages 3405.

An ESD transistor 3415 is coupled to one or more source follower stages3405 and may be configured to conduct a current greater than 500 mA whenexposed to an overvoltage pulse, as discussed below. Resistors 3410 aredisposed between source 3420 of ESD transistor 3415 and each source 3407of source follower stages 3405. Drains 3408 of source follower stages3405 are connected to drain 3425 of ESD transistor 3415. Source 3407 ofthe last source follower stage is coupled to gate 3430 of ESD transistor3415.

In one embodiment, a turn on voltage of ESD clamp circuit 3400 can beset by the total number of source follower stages 3405. However, sincethe last source follower stage is a transistor with a certain drain 3408to source 3407 voltage and gate 3406 to source voltage the currentthrough the final resistor 3410 may be relatively large and may resultin a larger gate 3430 to source 3420 voltage across ESD transistor 3415.This condition may result in a relatively large ESD current capabilityand in some embodiments an improved leakage performance compared toother ESD circuit configurations.

In further embodiments, ESD clamp circuit 3400 may have a plurality ofdegrees of freedom with regard to transistor sizes and resistor values.In some embodiments ESD clamp circuit 3400 may be able to be madesmaller than other ESD circuit configurations. In other embodiments, theperformance of ESD clamp circuit 3400 may be improved by incrementallyincreasing the size of source follower stages 3405 as they get closer toESD transistor 3415. In further embodiments, resistors 3410 can bereplaced by depletion-mode transistors, reference current sinks orreference current sources, for example.

Now referring to FIG. 35 an embodiment similar to ESD clamp circuit 3400in FIG. 34 is illustrated, however ESD clamp circuit 3500 may haveresistors in a different configuration, as discussed in more detailbelow. ESD clamp circuit 3500 may have a configuration employing one ormore source follower stages 3505 made from one or more enhancement-modetransistors. Each source follower stage 3505 may have a gate 3506connected to a source 3507 of an adjacent source follower stage. In theembodiment illustrated in FIG. 35, four source follower stages 3505 areemployed, however in other embodiments fewer or more may be used.Resistors 3510 are coupled between sources 3507 of adjacent sourcefollower stages 3505. An ESD transistor 3515 is coupled to sourcefollower stages 3505 with resistor 3510 disposed between source 3520 ofESD transistor 3515 and source 3507 of a source follower stage 3505.Drains 3508 of source follower stages 3505 may be coupled together andto drain 3525 of ESD transistor 3515.

Electronic Packaging

Now referring to FIGS. 36 and 37, in some embodiments, one or moresemiconductor devices may be disposed in one or more electronicpackages. Myriad packaging configurations and types of electronicpackages are available and are within the scope of this disclosure. FIG.36 illustrates one example of what is known as a quad-flat no-leadelectronic package with two semiconductor devices within it.

Electronic package 3600 may have a package base 3610 that has one ormore die pads 3615 surrounded by one or more terminals 3620. In someembodiments package base 3610 may comprise a leadframe while in otherembodiments it may comprise an organic printed circuit board, a ceramiccircuit or another material.

In the embodiment depicted in FIG. 36, a first device 3620 is mounted toa first die pad 3615 and a second device 3625 is mounted to a second diepad 3627. In another embodiment one or more of first and second devices3620, 3625, respectively may be mounted on an insulator (not shown) thatis mounted to package base 3610. In one embodiment the insulator may bea ceramic or other non-electrically conductive material. First andsecond devices 3620, 3625, respectively are electrically coupled toterminals 3640 with wire bonds 3630 or any other type of electricalinterconnect such as, for example, flip-chip bumps or columns that maybe used in a flip-chip application. Wirebonds 3630 may extend betweendevice bond pads 3635 to terminals 3640, and in some cases to die pads3615, 3627 and in other cases to device bond pads 3635 on an adjacentdevice.

Now referring to FIG. 37, an isometric view of electronic package 3600is shown. Terminals 3640 and die attach pads 3615 and 3627 may bedisposed on an external surface and configured to attach to a printedcircuit board or other device. In further embodiments, terminals 3640and die attach pads 3615 and 3627 may only be accessible within theinside of electronic package 3600 and other connections may be disposedon the outside of the electronic package. More specifically, someembodiments may have internal electrical routing and there may not be aone to one correlation between internal and external connections.

In further embodiments first and second devices 3620, 3625, respectively(see FIG. 36) and a top surface of package base 3610 may be encapsulatedby a non-electrically conductive material, such as for example, amolding compound. Myriad other electronic packages may be used such as,but not limited to, SOIC's, DIPS, MCM's and others. Further, in someembodiments each device may be in a separate electronic package whileother embodiments may have two or more electronic devices within asingle package. Other embodiments may have one or more passive deviceswithin one or more electronic packages.

In the foregoing specification, embodiments of the invention have beendescribed with reference to numerous specific details that may vary fromimplementation to implementation. The specification and drawings are,accordingly, to be regarded in an illustrative rather than a restrictivesense. The sole and exclusive indicator of the scope of the invention,and what is intended by the applicants to be the scope of the invention,is the literal and equivalent scope of the set of claims that issue fromthis application, in the specific form in which such claims issue,including any subsequent correction.

1. A level shift circuit comprising: a first inverter circuitcomprising: a first input terminal; a first output terminal; a firstGaN-based enhancement-mode transistor having a gate coupled to the firstinput terminal, a drain coupled to the first output terminal, and asource coupled to a ground; and a first pull up device coupled betweenthe drain of the first GaN-based enhancement-mode transistor and a powersupply; and a second inverter circuit comprising: a second inputterminal; a second output terminal; a second GaN-based enhancement-modetransistor having a gate coupled to the second input terminal, a draincoupled to the second output terminal, and a source coupled to theground; and a second pull up device coupled between the drain of thesecond GaN-based enhancement-mode transistor and the power supply,wherein the first pull up device comprises a third GaN-based transistorhaving a gate voltage which is different from a voltage at the firstoutput terminal, and wherein the first, second, and third transistorsare of the same conductivity type.
 2. The level shift circuit of claim 1wherein voltages generated at the first and the second input terminalsare referenced to a first voltage that is a ground, and voltagesgenerated at the first and the second output terminals are referenced toa second voltage at a different potential than ground.
 3. The levelshift circuit of claim 1, wherein the power supply is a floating powersupply.
 4. The level shift circuit of claim 1 having a first capacitancebetween the first output terminal and a floating voltage and a secondcapacitance between the first output terminal and ground, wherein thefirst capacitance is greater than the second capacitance.
 5. The levelshift circuit of claim 4 wherein an overvoltage condition on the firstoutput terminal is prevented by a clamp.
 6. The level shift circuit ofclaim 1 wherein the first inverter circuit input terminal is configuredto receive a first pulsed input signal from a first pulse generator andthe second inverter circuit input terminal is configured to receive asecond pulsed input signal from a second pulse generator.
 7. The levelshift circuit of claim 6 wherein at least one of the first pulsegenerator and the second pulse generators are configured to receiveinput pulses in a range of 2 nanoseconds to 20 microseconds and totransmit pulses of substantially constant duration within the range. 8.The level shift circuit of claim 6 wherein at least one of the firstpulse generator and the second pulse generators comprise at least onecombinatorial logic function.
 9. The level shift circuit of claim 6wherein the input signals from the first and the second pulse generatorscorrespond to on and off transitions of a pulse-width modulated (PWM)signal controlling a gate of a high side transistor.
 10. The level shiftcircuit of claim 6 further comprising a latching storage logic circuitconfigured to change state in response to a first pulsed input signalfrom the first pulse generator and to change state in response to asecond pulsed input signal from the second pulse generator.
 11. Thelevel shift circuit of claim 6 wherein the first and the second pulsedinput signals from the first and the second pulse generators,respectively, correspond to on and off transitions of a PWM signal tocontrol the gate of a high side transistor.
 12. The level shift circuitof claim 6 wherein at least one of the first and the second pulsegenerators are coupled with one or more logic gates.
 13. The level shiftcircuit of claim 1 further configured to generate a logical combinationof at least one PWM signal and at least one pulse generator outputsignal wherein the logical combination is used to prevent simultaneousconduction of a high side and a low side switch.
 14. The level shiftcircuit of claim 1 wherein an on level shift pulse can be shortened byan off input pulse to enable an on time of less than 50 nanoseconds on ahigh side switch.
 15. The level shift circuit of claim 1 wherein an offlevel shift pulse can be shortened by an on input pulse to enable an offtime of less than 50 nanoseconds on a high side switch.
 16. The levelshift circuit of claim 1 wherein the first output terminal is coupled toa circuit configured to charge a state storage capacitor referenced tothe second voltage.
 17. The level shift circuit of claim 1 wherein thesecond output terminal is coupled to a circuit configured to discharge astate storage capacitor that is referenced to the second voltage. 18.The level shift circuit of claim 1 wherein an output signal from one ofthe first or the second output terminals prevents a dv/dt induced changein a signal from the other of the first or the second output terminals.19. An electronic power conversion component comprising: a package base;one or more GaN-based dies secured to the package base and comprising: afirst inverter circuit comprising: a first input terminal; a firstoutput terminal; a first GaN-based enhancement-mode transistor having agate coupled to the first input terminal, a drain coupled to the firstoutput terminal, and a source coupled to a ground; and a first pull updevice coupled between the drain of the first GaN-based enhancement-modetransistor and a power supply; and a second inverter circuit comprising:a second input terminal; a second output terminal; a second GaN-basedenhancement-mode transistor having a gate coupled to the second inputterminal, a drain coupled to the second output terminal, and a sourcecoupled to the ground; and a second pull up device coupled between thedrain of the second GaN-based enhancement-mode transistor and the powersupply, wherein the first pull up device comprises a third GaN-basedtransistor having a gate voltage which is different from a voltage atthe first output terminal, and wherein the first, second, and thirdtransistors are of the same conductivity type.
 20. A method of operatingGaN-based level shift circuit, the method comprising: receiving one ormore control signals; generating a first pulse, the first pulseoperating a first inverter circuit configured to change a state of anoutput node, wherein the first pulse is generated based on one or morelogic levels of the one or more control signals; and generating a secondpulse, the second pulse operating a second inverter circuit configuredto change a state of the output node, wherein the second pulse isgenerated in response to a logic level transition of the one or morecontrol signals.
 21. A level shift circuit, comprising: a first invertercircuit comprising: a first input terminal; a first output terminal; afirst GaN-based enhancement-mode transistor having a gate coupled to thefirst input terminal, a drain coupled to the first output terminal, anda source coupled to a ground; and a first pull up device coupled betweenthe drain of the first GaN-based enhancement-mode transistor and a powersupply; a second inverter circuit comprising: a second input terminal; asecond output terminal; a second GaN-based enhancement-mode transistorhaving a gate coupled to the second input terminal, a drain coupled tothe second output terminal, and a source coupled to the ground; and asecond pull up device coupled between the drain of the second GaN-basedenhancement-mode transistor and the power supply; and a driver circuitconfigured to receive one or more control signals and to apply a firstsignal to the first input terminal of the first inverter and a secondsignal to the second input terminal of the second inverter, wherein thefirst signal is generated based on one or more logic levels of the oneor more control signals, and wherein the second signal is generated inresponse to a logic level transition of the one or more control signals.